ERIC-1006
`Ericsson v IV
`Page 1 of 12
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`U.S. Patent
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`Oct. 2, 2001
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`Sheet 1 0f5
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`US 6,298,035 B1
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`ERIC-1006 I Page 2 of 12
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`ERIC-1006 / Page 2 of 12
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`ERIC-1006 I Page 3 of 12
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`ERIC-1006 / Page 3 of 12
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`U.S. Patent
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`Oct. 2, 2001
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`Sheet 3 0f5
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`US 6,298,035 B1
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`ERIC-1006 I Page 4 of 12
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`U.S. Patent
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`Oct. 2, 2001
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`Sheet 4 0f5
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`US 6,298,035 B1
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`ERIC-1006 I Page 5 of 12
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`U.S. Patent
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`Oct. 2, 2001
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`Sheet 5 0f5
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`US 6,298,035 B1
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`
`
` 602
`Selecting appropriate training symbold
`for Channel 1 and Channel 2
`
` 604
`Transmitting training symbols and data
`from first and second transmitters
`
` 606
`Receiving transmitted training symbols
`and data at the receiver
`
`
`
`
`
`
` 608
`Combining first and second received
`symbols
`
` 610
`Estimating channel estimates for
`channels H1 and H2
`
`
`
`Figure 6
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`ERIC-1006 I Page 6 of 12
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`ERIC-1006 / Page 6 of 12
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`

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`US 6,298,035 B1
`
`1
`ESTIMATION OF TWO PROPAGATION
`CHANNELS IN OFDM
`
`FIELD OF THE INVENTION
`
`The invention relates to channel estimation, and more
`particularly to a method and apparatus for estimating two
`propagation channels in an Orthogonal Frequency Division
`Multiplexing (OFDM) system with two transmitter antennas
`using specifically selected training information.
`BACKGROUND OF THE INVENTION
`
`The growing area of personal communications systems is
`providing individuals with personal terminals capable of
`supporting various services such as multimedia. These ser-
`vices require the use of increased bit rates due to the large
`amount of data required to be transferred. The use of
`increased bit rates generates problems in conventional single
`carrier systems due to inter-symbol interference (ISI) and
`deep frequency selective fading problems.
`One solution to these problems utilizes orthogonal fre-
`quency division multiplexing (OFDM) within the radio
`mobile environment to minimize the above-mentioned prob-
`lems. Within OFDM, a signal
`is transmitted on multi-
`orthogonal carriers having less bandwidth than the coher-
`ence bandwidth of the channel in order to combat frequency
`selective fading of the transmitted signal. The inter-symbol
`interference is mitigated by the use of guard intervals.
`OFDM systems are presently adopted in Europe for digital
`audio broadcasting and have been proposed for use in digital
`TV broadcasting systems. It is used also in asymmetric
`digital subscriber lines (ADSL) to transmit high rate data.
`OFDM has also been selected as the modulation method for
`
`wireless local area network (WLAN) standards in United
`States, Europe and Japan.
`Transmitter diversity is an effective technique to mitigate
`multipath fading. One significant advantage of transmitter
`diversity is that the receiver needs only one antenna with
`Radio Frequency (RF) receiving chain. Since RF compo-
`nents are quite expensive the cost of the receiver can be
`reduced with transmitter diversity compared to a system
`using receiver diversity, that needs two or more antennas and
`corresponding receiving RF chains. Recently Space-Time
`Codes (STC) have been introduced as a method to achieve
`transmitter diversity system. Space-Time codes encode
`information over multiple antennas to achieve diversity
`advantage, however decoding of STC needs an estimate of
`the propagation path from each transmitter antenna to the
`receiver antenna.
`
`Since radio channels often are subjected to multipath
`propagation, the receiver needs to comprise some sort of
`equalizer to eliminate this phenomenon. The equalizer
`requires an estimated frequency response of the transmission
`channel, i.e., a channel estimation. Existing channel estima-
`tion methods are based on adaptive signal processing
`wherein the channels are assumed to vary slowly. The
`estimated channel parameters at a particular time depend on
`the received data and channel parameters at a previous time.
`In the case of fast varying channels, such as in high data rate
`mobile systems, these methods must be modified to reduce
`the estimation time.
`
`Single channel estimation is a well known problem and
`numerous methods exist to solve that problem. However,
`their extension to estimating multiple channels in an OFDM
`system has not been discussed. For example, Space-Time
`coded communication systems use multiple transmit anten-
`nas to achieve transmitter diversity gain, but require each
`
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`2
`propagation channel to be separately estimated. A trivial
`way to use existing single channel estimation algorithms is
`to separate transmission of the training information in time
`for each transmit antenna. Then, the existing algorithms can
`be used for each antenna as each antenna is transmitting
`training information.
`A drawback of separating the training information in time
`is that it reduces the amount of information used to estimate
`each channel, provided that a fixed amount of training data
`is available. Time divisioning the training data between two
`antennas decreases the quality of the estimate of each
`channel. Another option is to double the amount of training
`data, which in turn increases the system overhead.
`SUMMARY OF THE INVENTION
`
`It is an object of the present invention to overcome the
`deficiencies described above by providing a method and
`apparatus for estimating separate channel frequency
`responses in a communication system with two transmitters.
`The channel frequency responses are estimated using spe-
`cifically selected training symbols that are broadcast from
`the two transmitters. The invention has the advantage of
`retaining the same amount of training symbols as required in
`a single channel estimation case while improving the chan-
`nel estimate for each channel.
`
`According to one embodiment of the present invention, a
`method and apparatus for estimating separate channel fre-
`quency responses for channels in an orthogonal frequency
`division multiplexing system with two transmitters is dis-
`closed. First and second training symbols (A1, A2) and data
`from a first transmitter are transmitted to a receiver using a
`first channel. Third and fourth training symbols (B1, B2) and
`data from a second transmitter are transmitted to the receiver
`
`using a second channel. First and second received symbols
`are received at the receiver. The first and second received
`
`symbols are then combined. A first channel estimate and a
`second channel estimate are then derived from the combined
`
`received symbols, wherein the first and second received
`symbols comprise the training symbols, wherein the first and
`third training symbols form a first symbol pair and the
`second and fourth training symbols form a second symbol
`pair.
`
`BRIEF DESCRIPTION OF THE FIGURES
`
`For a better understanding of these and other objects of
`the present
`invention, reference is made to the detailed
`description of the invention, by way of example, which is to
`be read in conjunction with the following drawings,
`wherein:
`
`FIG. 1 is a block diagram of a typical OFDM transmitter
`according to the prior art;
`FIG. 2 is an illustration of a typical OFDM signal within
`an OFDM channel bandwidth showing the frequency
`domain positioning of OFDM sub-carriers and their modu-
`lated spectra, according to the prior art;
`FIG. 3 is a block diagram of a typical OFDM receiver
`according to the prior art;
`FIG. 4 is a block diagram of an OFDM communication
`system with two transmit antennas and one receive antenna
`according to one embodiment of the invention;
`FIG. 5 illustrates the transmission of training information
`according to one embodiment of the invention; and
`FIG. 6 is a flow chart illustrating a channel estimation
`process according to one embodiment of the invention.
`DETAILED DESCRIPTION
`
`Orthogonal frequency division multiplexing is a robust
`technique for efficiently transmitting data over a channel.
`
`ERIC-1006 I Page 7 of 12
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`US 6,298,035 B1
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`3
`The technique uses a plurality of sub-carrier frequencies
`(sub-carriers) within a channel bandwidth to transmit the
`data. These sub-carriers are arranged for optimal bandwidth
`efficiency compared to more conventional
`transmission
`approaches, such as frequency division multiplexing
`(FDM), which waste large portions of the channel band-
`width in order to separate and isolate the sub-carrier fre-
`quency spectra and thereby avoid intercarrier interference
`(ICI). By contrast, although the frequency spectra of OFDM
`sub-carriers overlap significantly within the OFDM channel
`bandwidth, OFDM nonetheless allows resolution and recov-
`ery of the information that has been modulated onto each
`sub-carrier. Additionally, OFDM is much less susceptible to
`data loss due to multipath fading than other conventional
`approaches for data transmission because intersymbol inter-
`ference is prevented through the use of OFDM symbols that
`are long in comparison to the length of the channel impulse
`response. Also,
`the coding of data onto the OFDM sub-
`carriers can take advantage of frequency diversity to miti-
`gate loss due to frequency-selective fading.
`
`The general principles of OFDM signal transmission can
`be described with reference to FIG. 1 which is a block
`
`diagram of a typical OFDM transmitter according to the
`prior art. An OFDM transmitter 10 receives a stream of
`baseband data bits 12 as its input. These input data bits 12
`are immediately fed into an encoder 14, which takes these
`data bits 12in segments of B bits every Tg+T5 seconds,
`where T5 is an OFDM symbol interval and T8, is a cyclic
`prefix or guard interval. The encoder 14 typically uses a
`block and/or convolutional coding scheme to introduce
`error-correcting and/or error-detecting redundancy into the
`segment of B bits and then sub-divides the coded bits into
`2N sub-segments of m bits. The integer m typically ranges
`from 2 to 6.
`
`there are 2N+1 OFDM
`In a typical OFDM system,
`sub-carriers, including the zero frequency DC sub-carrier
`which is not generally used to transmit data since it has no
`frequency and therefore no phase. Accordingly, the encoder
`14 then typically performs 2'”-ary quadrature amplitude
`modulation (QAM) encoding of the 2N sub-segments of m
`bits in order to map the sub-segments of m bits to prede-
`termined corresponding complex-valued points in a 2'”-ary
`constellation. Each complex-valued point in the constella-
`tion represents discrete values of phase and amplitude. In
`this way, the encoder 14 assigns to each of the 2N sub-
`segments of m bits a corresponding complex-valued 2'”-ary
`QAM sub-symbol ck=ak+jbk, where —N1 §k§N1, in order to
`create a sequence of frequency-domain sub-symbols that
`encodes the B data bits. Also, the zero-frequency sub-carrier
`is typically assigned cO=0. The encoder 14 then passes the
`sequence of subsymbols, along with any additional zeros
`that may be required for interpolation to simplify filtering,
`on to an inverse discrete Fourier transformer (IDFT) or,
`preferably, an inverse fast Fourier transformer (IFFT) 16.
`
`Upon receiving the sequence of OFDM frequency-
`domain sub-symbols from the encoder 14,
`the IFFT 16
`performs an inverse Fourier transform on the sequence of
`sub-symbols. In other words, it uses each of the complex-
`valued sub-symbols, ck, to modulate the phase and ampli-
`tude of a corresponding one of 2N+1 sub-carrier frequencies
`over a symbol interval T5. The sub-carriers are given by
`e‘2“’7k’, and therefore, have baseband frequencies of fk=k/T5,
`where k is the frequency number and is an integer in the
`
`4
`range —N§k§ N. The IFFT 16 thereby produces a digital
`time-domain OFDM symbol of duration T5 given by
`N
`
`k:—N
`um = Z ckexp(—27rfikt).
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`As a result of this discrete-valued modulation of the
`
`OFDM sub-carriers by frequency-domain sub-symbol inter-
`vals of T5 seconds, the OFDM sub-carriers each display a
`sinc x=(sin x)/x spectrum in the frequency domain. By
`spacing each of the 2N+1 sub-carriers 1/T5 apart in the
`frequency domain, the primary peak of each sub-carriers
`sinc x spectrum coincides with a null of the spectrum of
`every other sub-carrier. In this way, although the spectra of
`the sub-carriers overlap,
`they remain orthogonal
`to one
`another. FIG. 2 illustrates the arrangement of the OFDM
`sub-carriers as well as the envelope of their modulated
`spectra within an OFDM channel bandwidth, BW, centered
`around a carrier frequency, fa. Note that the modulated
`sub-carriers fill the channel bandwidth very efficiently.
`Returning to FIG. 1,
`the digital
`time-domain OFDM
`symbols produced by the IFFT 16 are then passed to a digital
`signal processor (DSP) 18. The DSP 18 performs additional
`spectral shaping on the digital time-domain OFDM symbols
`and also adds a cyclic prefix or guard interval of length T8,
`to each symbol. The cyclic prefix is generally just a repeti-
`tion of part of the symbol. This cyclic prefix is typically
`longer than the OFDM channel
`impulse response and,
`therefore, acts to prevent inter-symbol
`interference (ISI)
`between consecutive symbols.
`The real and imaginary-valued digital components that
`make up the cyclically extended, spectrally-shaped digital
`time-domain OFDM symbols are then passed to digital-to-
`analog converters (DACs) 20 and 22, respectively. The
`DACs 20 and 22 convert the real and imaginary-valued
`digital components of the time-domain OFDM symbols into
`in-phase and quadrature OFDM analog signals, respectively,
`at a conversion or sampling rate of fckir as determined by a
`clock circuit 24. The in-phase and quadrature OFDM signals
`are then passed to mixers 26 and 28, respectively.
`In the mixers 26 and 28,
`the in-phase and quadrature
`OFDM signals from the DACs 20 and 22 are used to
`modulate an in-phase intermediate frequency signal (IF) and
`a 90° phase-shifted (quadrature) IF signal, respectively, in
`order to produce an in-phase IF OFDM signal and a quadra-
`ture IF OFDM signal, respectively. The in-phase IF signal
`that is fed to the mixer 26 is produced directly by a local
`oscillator 30, while the 90° phase-shifted IF signal that is fed
`to the mixer 28 is produced by passing the in-phase IF signal
`produced by the local oscillator 30 through a 90° phase-
`shifter 32 before feeding it
`to the mixer 28. These two
`in-phase and quadrature IF OFDM signals are then com-
`bined in a combiner 34 to form a composite IF OFDM
`signal.
`In some prior art
`transmitters,
`the IF mixing is
`performed in the digital domain using a digital synthesizer
`and digital mixers before the digital-to-analog conversion is
`performed.
`This composite IF OFDM signal is then passed into radio
`frequency transmitter 40. Many variations of RF transmitter
`40 exist and are well known in the art, but typically, the RF
`transmitter 40 includes an IF bandpass filter 42, an RF mixer
`44, an RF carrier frequency local oscillator 46, an RF
`baseband filter 48, an RF power amplifier 50, and an antenna
`52. The RF transmitter 40 takes the IF OFDM signal from
`the combiner 34 and uses it to modulate a transmit carrier of
`
`ERIC-1006 I Page 8 of 12
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`

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`US 6,298,035 B1
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`5
`frequency fa, generated by the RF local oscillator 46, in
`order to produce an RF OFDM-modulated carrier that
`occupies a channel bandwidth, BM. Because the entire
`OFDM signal must fit within this channel bandwidth, the
`channel bandwidth must be at least (1/T5)~(2N+1) Hz wide
`to accommodate all the modulated OFDM sub-carriers. The
`frequency-domain characteristics of this RF OFDM-
`modulated carrier are illustrated in FIG. 2. This RF OFDM-
`modulated carrier is then transmitted from antenna 52
`
`to an OFDM receiver in a remote
`through a channel,
`location. In alternative embodiments of RF transmitters, the
`OFDM signal is used to modulate the transmit carrier using
`frequency modulation, single-sided modulation, or other
`modulation techniques. Therefore, the resulting RF OFDM-
`modulated carrier may not necessarily have the exact shape
`of the RF OFDM-modulated carrier illustrated in FIG. 2, i.e.,
`the RF OFDM-modulated carrier might not be centered
`around the transmit carrier, but instead may lie to either side
`of it.
`
`In order to receive the OFDM signal and to recover at a
`remote location the baseband data bits that have been
`encoded into the OFDM sub-carriers, an OFDM receiver
`must perform essentially the inverse of all of the operations
`performed by the OFDM transmitter described above. These
`operations can be described with reference to FIG. 3 which
`is a block diagram of a typical OFDM receiver according to
`the prior art.
`The received signal is first filtered in a receiver filter 302
`so as to limit the bandwidth of the received signal. The band
`limited received signal is then sent to a channel estimator
`304, wherein the channel estimator comprises a processor.
`The channel estimator processes the band limited received
`signal
`to produce an estimate of the channel frequency
`response
`Of the transmit channel. In this example, the
`channel estimator also performs frame synchronization in a
`known manner and produces an estimate of the frame timing
`(Ta.
`A
`The estimate of the frame timing (TF) is sent to S/P
`processor 306 which converts the serial data input stream
`from the receive filter 302 and frame timing from the
`channel estimator into a parallel stream by framing N
`symbols. The S/P 306 outputs a received cyclically extended
`OFDM frame. The cyclic prefix attached to the OFDM data
`frame is then removed in processor 308. With proper
`synchronization,
`the inter-frame interference is removed.
`The received OFDM data frame is then sent to the Discrete
`
`Fourier Transformer DFT 310. The DFT 310 implements the
`OFDM demodulator with N sub-carriers using the discrete
`Fourier transform, wherein the input corresponds to the time
`domain and the output to the frequency domain. The DFT
`310 outputs the transmitted modulated symbols affected by
`the channel frequency response to a channel equalizer 312.
`The channel equalizer 312 receives the estimated channel
`frequency response and the transmitted modulated signals.
`The channel equalizer 312 performs frequency domain
`zero-forcing equalization of the OFDM sub-carriers. Only
`sub-carriers with magnitudes above a certain predetermined
`threshold value are equalized, since magnitudes below the
`predetermined threshold value are considered unreliable.
`The channel equalizer 312 outputs recovered modulated
`signals. The recovered modulated signals are converted
`from N-symbol parallel data streams (frames) into a serial
`stream in a P/S processor 314. The serial stream is then
`inputted into a base band demodulator 316. The base band
`demodulator 316 demodulates the recovered modulated sig-
`nals and maps one input symbol into k binary symbols
`according to the base band signaling scheme. The base band
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`demodulator outputs received binary data to a data sink 318
`which applies application specific processing to the received
`data.
`
`FIG. 4 shows a model of an OFDM communication
`
`system 400 with two transmit antennas and one receive
`antenna. This system has two separate propagation channels
`H1 and H2. The goal of this embodiment of the present
`invention is to estimate the channel frequency response of
`both of these channels using the structure of the training
`information. A first transmitter 402 prepares information to
`be transmitted, for example in the manner set forth above
`with respect to FIG. 1, and the information is sent to a
`transmit filter 404 and then transmitted to the receiver 414
`
`through the physical channel H1 (406). During transmission,
`noise is unavoidably added to the transmitted signal. A
`second transmitter 408 prepares information to be transmit-
`ted and sends the information to a transmit filter 410. The
`
`information is then transmitted to the receiver 414 through
`the physical channel H2 (412). During transmission, noise is
`unavoidably added to the transmitted signal. When the
`signals are received at
`the receiver 414,
`the signals are
`filtered in a receive filter 416 and are then processed in a
`processor 418. One of the operations of the processor 418 is
`to estimate a channel frequency response of channels H1 and
`H2.
`
`FIG. 5 shows how the training information is transmitted
`by the two transmitters 402 and 408. The first transmitter
`402 transmits OFDM training symbols A1 and A2, and the
`second transmitter 408 transmits OFDM training symbols
`B1 and B2. The goal of the receiver is to separate the OFDM
`symbols so that all the information in A1 and A2 can be used
`to estimate the channel frequency response of channel H1
`and all of the information in B1 and B2 can be used to
`
`estimate the channel frequency response of channel H2.
`
`The operation of one embodiment of the present invention
`will now be described with reference to FIG. 6. As will be
`
`explained below in more detail, the transmitters 402 and 408
`select
`the appropriate training symbols in step 602 and
`transmit the training symbols and data over physical chan-
`nels H1 and H2, respectively, in step 604. The transmitted
`training symbols and data are then received at the receiver
`414 in step 606. The first received symbol R1 at the receiver
`in frequency-domain during the transmission of the training
`symbols A1 and B1, with additive noise N1 is
`
`R1=H1-A1+H2-B1+N1
`
`and the second received symbol R2 during transmission of
`the training symbols A2 and B2, with additive noise N2 is
`
`R2=H1-A2+H2-B2+N2
`
`To achieve noise reduction, the signals R1 and R2 are added
`together in step 608
`
`R=R1+R2=H1-A1+H2-B1+H1-A2+H2-B2+N1+N2
`
`After reordering the terms
`
`R=H1-(A1+A2)+H2-(B1+B2)+N1+N2.
`
`To estimate H1 it is necessary to remove the effects of H2
`on the received signal R and vice versa. As a result, B1+B2
`should be equal to zero while preserving A1+A2, and vice
`versa. One solution according to one embodiment of the
`
`ERIC-1006 I Page 9 of 12
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`ERIC-1006 / Page 9 of 12
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`

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`US 6,298,035 B1
`
`7
`present invention is to select A1, A2, B1 and B2 as follows:
`A1 is a set of complex numbers, one number for each
`subcarrier
`
`A2=A1
`
`B1=A1
`
`B2=A1
`
`|A1|2=1
`
`and
`
`In this case, the sum of R1 and R2 is
`R=R1+R2=H1-A1+H2-A1+H1-A1—H2-A1+N1+N2=2-H1-A1+
`H2-(A1—A1)+N1+N2=2-H1-A1+N1+N2
`
`In step 610, the channel frequency response A1 can now
`be estimated by multiplying R by
`
`A1*-R+N1+N2
`2
`
`H1:
`
`2 N1+N2
`=H1-|A1| +
`2
`
`=H1+
`
`N1+N2
`2
`
`A1 conjugate and dividing by 2. Since noise is independent,
`its power is reduced by 2.
`Similarly, the channel frequency response H2 of channel
`H2 can be estimated by subtracting R1 and R2.
`
`R=R1—R2+N1+N2=H1-A1+H2-A1—H1-A1+H2-A1+N1—N2=
`H1-(A1—A2)+2H2-A1+N1—N2+2-H2-A1+N1—N2
`
`Now, the channel frequency response H2 can be estimated
`by multiplying R by A1 conjugate and
`
`1v1—1v2_H2
`H2_A1*-R_H2 A1,
`‘
`2
`‘
`"
`'+ 2
`‘
`A1*-R
`2
`1v1—1v2
`2
`=H2-|A1| +
`2
`
`2
`H2:
`
`=H2+
`
`J"
`
`1v1—1v2
`2
`1v1—1v2
`2
`
`dividing by 2.
`One drawback with this solution is, if H1=H2, as in the
`additive White Gaussian Noise (AWGN) channel,
`the
`received signal during A2 and B2 is equal to A1—A1=0, so
`nothing is received. To remove this effect, the symbol pairs
`(A1,B1) and (A2,B2) should be orthogonal. In this case,
`they will not cancel each other, if the channels H1 and H2
`happen to be equal.
`According to another embodiment of the present
`invention, the following selection of the symbols A1, A2,
`B1, B2 has all of the required properties to avoid the
`problems in the additive White Gaussian Noise channel.
`Symbol pairs (A1, B1) and (A2, B2) have a 90° phase-shift,
`so they are orthogonal and will not cancel each other out in
`an Additive Gaussian Noise channel. Also,
`the channel
`estimation can be performed using both A1 and A2 for H1,
`and B1 and B2 for H2.
`
`A1 is a set of complex numbers, one number for each
`subcarrier
`
`A2=A1
`
`B1=e/"/2A1
`
`B2=e*f"/2A1
`
`and
`
`|A1|2=1
`
`8
`With these training symbols, the channel estimation can be
`performed in the following manner for H1:
`R=R] +R2+N]+N2=H] -A]+H2-ef"/2-A1+H1-A1+H2-e’j"/2-A1+N1+
`N=2-H1-A1+H2-A1(e/"/2+e’j"/2)+N1+N2=2-H1-A1+N1+N2
`
`Now, the channel frequency response H1 can be estimated
`by multiplying R by A1 conjugate and
`
`H1:
`
`A1*-R
`2
`
`2
`=H1-|A1| +
`
`N1+N2 _
`2
`_H1+
`
`N1+N2
`2
`
`dividing by two:
`The estimation of the channel frequency response of
`channel H2 can then be performed as follows:
`
`R=e*f"/2-R1+eJ"/2-R2=H1-e*f"/2-A1+H2-A1+H1-ei"/2-A1+H2-A1+
`N1 +N2=H1-A1-(e’/"/2)+2-H2-A1+N1+N2=2-H2-A1+N1+N2
`
`10
`
`15
`
`20
`
`Now, the channel frequency response of H2 can be estimated
`with the following equation:
`
`2 N1+N2
`=H2-|A1| +
`H2 A1*-R+N1+N2W 2
`2
`
`=H2+
`
`N1+N2
`2
`
`Although preferred embodiments of the method and appa-
`ratus of the present invention have been illustrated in the
`accompanying Drawings and described in the foregoing
`Detailed Description, it is understood that the invention is
`not limited to the embodiments disclosed, but is capable of
`numerous rearrangements, modifications, and substitutions
`without departing from the spirit or scope of the invention as
`set forth and defined by the following claims.
`What is claimed is:
`
`1. A method for estimating separate channel frequency
`responses for channels in an orthogonal frequency division
`multiplexing system with two transmitters, each having an
`antenna, comprising the steps of:
`selecting training symbols for each said antenna that
`allow for separately estimating the frequency response
`of each channel;
`transmitting the training symbols selected for a first
`antenna from a first transmitter to a receiver using a first
`channeh
`transmitting the training symbols selected for a second
`antenna from a second transmitter to the receiver using
`a second channel;
`receiving training symbols at the receiver; and
`estimating a first channel estimate and a second channel
`estimate from the received training symbols.
`2. The method according to claim 1, wherein said symbols
`are orthogonal.
`3. The method according to claim 1, wherein A1 and A2
`are training symbols for the first antenna, B1 and B2 are
`training symbols for the second antenna, wherein
`A1 is a set of complex numbers, one number for each
`subcarrier
`
`25
`
`30
`
`35
`
`40
`
`45
`
`50
`
`55
`
`60
`
`and
`
`65
`
`A2=A1
`
`B1=A1
`
`B2=—A1
`
`|A1|2=1.
`
`ERIC-1006 I Page 10 of 12
`
`ERIC-1006 / Page 10 of 12
`
`

`
`US 6,298,035 B1
`
`9
`4. The method according to claim 1, wherein a first
`received training symbol (R1) equals
`R1=H1-A1+H2-B1+N1
`
`and a second received training symbol R2 equals
`R2=H1-A2+H2-B2+N2
`
`10
`A1 is a set of complex numbers, one number for each
`subcarrier
`
`A2=A1
`
`B1=A1
`
`B2=—A1
`
`wherein N1 and N2 are noise.
`
`and
`
`5. The method according to claim 3, wherein the first 10
`channel estimate (H1) equals
`
`|A1|2=1.
`
`H1=H1+(N1+N2 )/2
`
`and the second channel estimate (H2) equals
`
`15
`
`H2=H2+(N1+N2)/2
`
`wherein N1 and N2 are noise.
`
`6. The method according to claim 1, wherein A1 and A2
`are training symbols for the first antenna, B1 and B2 are
`training symbols for the second antenna, wherein
`A1 is a set of complex numbers, one number for each
`subcarrier
`
`10. The system according to claim 8, wherein the first
`received symbol (R1) equals
`R1=H1-A1+H2-B1+N1
`
`and the second received symbol R2 equals
`R2=H1-A2+H2-B2+N2
`
`wherein N1 and N2 are noise.
`
`11. The system according to claim 9, wherein the first
`channel estimate (H1) equals
`
`H1=H1+(N1+N2)/2
`
`and the second channel estimate (H2) equals
`
`H2=H2+(N1+N2)/2
`
`wherein N1 and N2 are noise.
`
`25
`
`30
`
`A2=A1
`
`B1=e/"/2A1
`
`B2=e*f"/2A1
`
`and
`
`|A1|2=1.
`
`7. The method according to claim 6, wherein the first
`channel estimate (H1) equals
`
`35
`
`12. The system according to claim 8, wherein A1 and A2
`are training symbols for the first antenna, B1 and B2 are
`training symbols for the second antenna
`A1 is a set of complex numbers, one number for each
`subcarrier
`
`A2=A1
`
`B1=e/"/2A1
`
`B2=e*f"/2A1
`
`and
`
`|A1|2=1.
`
`13. The system according to claim 12, wherein the first
`channel estimate (H1) equals
`
`H1=H1+(N1+N2)/2
`
`40
`
`45
`
`50
`
`H1=H1+(N1+N2)/2
`
`and the second channel estimate (H2) equals
`
`H2=H2+(N1+N2)/2
`
`wherein N1 and N2 are noise.
`
`8. A system for estimating separate channel frequency
`responses for channels in an orthogonal frequency division
`multiplexing system with two transmitter antennas, com-
`prising:
`training symbols selected for each antenna that allow for
`separately estimating the frequency response of each
`channel;
`transmits the training symbols
`a first
`transmitter that
`selected for the first antenna and data from a first
`
`transmitter to a receiver using a first channel;
`a second transmitter that transmits the training symbols
`selected for the second antenna and data from a second
`
`transmitter to the receiver using a second channel;
`the receiver which receives received training symbols at
`the receiver;
`a combiner that combines said first and second received
`
`symbols; and
`a processor that estimates a first channel estimate and a
`second channel estimate from the combined received
`
`symbols.
`9. The system according to claim 8, wherein A1 and A2
`are training symbols for the first antenna, B1 and B2 are
`training symbols for the second antenna, wherein
`
`and the second channel estimate (H2) equals
`
`H2=H2+(N1+N2)/2
`
`55
`
`wherein N1 and N2 are noise.
`
`14. A receiver for receiving signals from two transmitters
`over two channels, comprising:
`a filter for filtering first (R1) and second (R2) received
`symbols transmitted from first and second transmitters,
`wherein said first received symbol comprises a first
`training symbol (A1) from the first transmitter and a
`first training symbol (B1) from a second transmitter and
`the second received symbol (R2) comprises a second
`training symbol
`from the first transmitter and a
`second training symbol
`(B2)
`from the second
`transmitter, said training symbols being affected by
`each channel’s frequency response;
`
`60
`
`65
`
`ERIC-1006 I Page 11 of 12
`
`ERIC-1006 / Page 11 of 12
`
`

`
`US 6,298,035 B1
`
`12
`
`18. The receiverAaccording to claim 16, wherein the first
`channel estimate (H1) equals
`
`H1=H1+(N+N2)/2
`
`and the second channel estimate (H2) equals
`
`H2=H2+(N1+N2)/2
`
`wherein N1 and N2 are noise.
`
`19. The receiver according to claim 14, wherein
`A1 is a set of complex numbers, one number for each
`subcarrier
`
`11
`a combiner for combining the first and second received
`symbols; and
`
`a processor for estimating a first channel estimate and a
`second channel estimate from the combined received
`
`symbols.
`15. The receiver according to claim 14, wherein said
`symbols are orthogonal.
`16. The receiver according to claim 14, wherein
`
`A1 is a set of complex numbers, one number for each subcarrier
`
`A2=A1
`
`B1=A1
`
`B2=—A1
`
`|A1|2=1.
`
`and
`
`5
`
`10
`
`15
`
`A2=A1
`
`B1=e/"‘/2A1
`
`B2=e*f"/2A1
`
`|A1|2=1.
`
`and
`
`20
`
`17. The receiver according to claim 14, wherein the first
`received symbol (R1) equals
`
`25
`
`R1=H1-A1+H2-B1+N1
`
`20. The receiver according to claim 19, wherein the first
`channel estimate (H1) equals
`
`H1=H1+(N1+N2)/2
`
`and the second channel estimate (H2) equals
`
`and the second received symbol R2 equals
`
`R2=H1-A2+H2-B2+N2
`
`wherein N1 and N2 are noise.
`
`30
`
`H2=H2+(N1+N2)/2
`
`wherein N1 and N2 are noise.
`
`ERIC-1006 I Page 12 of 12
`
`ERIC-1006 / Page 12 of 12