Ulllted States Patent [19]
`Letaief et al.
`
`US005960032A
`[11] Patent Number:
`[45] Date of Patent:
`
`5,960,032
`Sep. 28, 1999
`
`[54] HIGH SPEED DATA TRANSMISSION USING
`EXPANDED BIT DURATIONS IN MULTIPLE
`PARALLEL CODED DATA STREAMS
`
`[75] Inventors: Khaled Ben Letaief, Taikoo Shing;
`Justin C_I Chuang; ROSS D_ Murch,
`both of Clearwater Bay, an of The
`Hong Kong Special Administrative
`
`8/1993 Murai .................................... .. 375/267
`5,239,541
`5,373,502 12/1994 Turban ......... ..
`.. 370/441
`5,414,734
`5/1995 Marchetto et a1
`375/267
`5,442,625
`8/1995 Gitlin et a1.
`370/342
`375/206
`5,467,367 11/ 1995 IZllIIli et a1
`5,521,937
`5/1996 Kondo et a1.
`375/206
`5,544,167
`8/1996 Lucas et a1.
`370/342
`5,555,268
`9/1996 Fattouche et a1. .................... .. 375/206
`
`
`
`Region of the peoplezs Republic of China
`
`
`
`Padovani et a1. ..................... .. 5,583,851 12/1996 Kato et a1. ........................ .. 375/206
`
`[73] Assignee: The Hong Kong University of Science
`& Tgchnolqgy’ The. Hong Kong speflal
`Administrative Region of the People s
`Republic of China
`
`5,615,227
`
`3/1997 Schumacher, Jr. et a1. .......... .. 375/206
`
`Primary Examiner—Don N. V0
`k
`.
`Attorney, Agent, or Ftrm—Burns, Doane, SWec er &
`Mathis, LLP
`
`[21] Appl. No.: 08/531,012
`
`[57]
`
`ABSTRACT
`
`Sep' 20’ 1995
`[22] Filed:
`[51] Int. Cl? ................................................... .. H04B 1/707
`[52] US. Cl. ............................................................ .. 375/206
`[58] Field of Search
`375/206 260
`335’ 342’
`’ 441’, 479’
`
`’
`
`’
`
`’
`
`High-rate bit transmitted data is serial to parallel converted
`into low-rate bit stream in a Similar fashion to multicarrier
`of multitone modulation Howe/V691“ Contrast ‘0 the mul
`ticarrier method, each loW-rate bit stream is modulated using
`direct-sequence spread-spectrum. By selecting the process
`ing gain properly the total required bandwidth Will be of the
`same order as the original high-rate data stream, thereby
`ainin the inherent bene?t of multi ath re'ection Without
`g
`g
`P
`J
`expanding the bandwidth of the original high-rate stream.
`
`56
`[
`]
`
`References Cited
`
`U.S. PATENT DOCUMENTS
`
`3,518,547
`
`6/1970 Filipowsky ........................... .. 370/208
`
`15 Claims, 7 Drawing Sheets
`
`Symbol duration = K Tb
`
`1
`
`b1 (0
`
`E
`5
`Data
`w
`—> ‘3
`Tb
`%
`6
`w
`
`r(t)
`
`?cosmct + w)
`
`ERIC-1001
`Ericsson v. IV
`Page 1 of 15
`
`

`
`U.S. Patent
`
`Sep.28, 1999
`
`Sheet 1 of7
`
`5,960,032
`
`Symbol duration = K Tb
`
`l
`
`Serial to Parallel
`
`1/500“ wot + w)
`
`"('0
`
`TH‘
`.(-)Jdt
`II]
`
`zj
`t__ T
`
`output A
`ill
`b (t)
`k
`data
`
`ak? - 7]) coskvct‘wj)
`
`FIG. 1
`
`Page 2 of 15
`
`

`
`U.S. Patent
`
`Sep.28, 1999
`
`Sheet 2 of7
`
`5,960,032
`
`10
`
`
`
`Error Probability
`
`10;
`
`10-35
`
`1°
`
`:
`
`—— Walsh xPN Code
`-------- -- Walsh Codes
`----- PN Sequences
`
`10-5
`
`l
`
`I ‘'"III
`
`'
`
`' ""lll
`
`I
`
`l lllllll
`
`.0001
`
`.1
`.01
`.001
`Normalized Delay Spread d
`
`1
`
`FIG. 2
`
`Page 3 of 15
`
`

`
`Sep.28, 1999
`
`Sheet 3 of 7
`
`5,960,032
`
`>a
`."'.:'
`
`'5
`N
`.D
`OL.
`D.
`1..
`Cx.5..
`UJ
`
`:- Walsh x PN Code
`
`-------- -- Walsh Codes
`----- PN Sequences
`
`1
`
`Resolution r
`
`FIG. 3
`
`Page 4 of 15
`
`

`
`U.S. Patent
`
`Sep.28, 1999
`
`Sheet 4 0f 7
`
`5,960,032
`
`102:
`
`-
`
`K=64
`
`£33153;
`
`:
`
`5
`(U
`.Q
`9
`D_
`é
`LE
`
`..
`
`.
`.
`.
`
`- 104;
`
`:
`
`K=32
`
`10's
`
`1
`
`. Inn-l .
`
`. lunn .
`
`. |H|H|
`
`I
`
`. llllll
`
`.0001
`
`.1
`.01
`.001
`Normalized Delay Spread d
`
`1
`
`FIG. 4
`
`Page 5 of 15
`
`

`
`U.S. Patent
`
`Sep.28, 1999
`
`Sheet 5 of7
`
`5,960,032
`
`10‘2_
`
`§
`L1
`‘5
`£163."
`
`.
`
`.
`-
`
`K=100
`
`_Q
`(U
`.Q
`9
`a
`
`-
`.
`
`K=64
`
`1 llllllll
`
`| |||||||l
`
`||||||||l
`
`| |||||||
`
`.0001
`
`1
`.01
`.001
`Normalized Delay Spread d
`
`1
`
`FIG. 5
`
`Page 6 of 15
`
`

`
`U.S. Patent
`
`Sep.28, 1999
`
`Sheet 6 of7
`
`5,960,032
`
`
`
`@5821 6am
`
`10'4
`.001
`
`1
`
`Normalized Delay Spread d
`
`FIG. 6
`
`Page 7 of 15
`
`

`
`U.S. Patent
`
`Sep.28, 1999
`
`Sheet 7 0f 7
`
`5,960,032
`
`Zn
`
`|——_' Decision —> who]
`
`(i=0)
`Tb lfn_1[o]., ............................................................................. .=
`
`Zn_1
`
`- Zn-1[o]
`
`Decision
`
`A_
`»’b(,'1)_1[1]
`
`v
`
`Decision —> 8922p]
`
`Decision —> 352mm]
`
`Channel Estimation
`
`FIG. 7
`
`Page 8 of 15
`
`

`
`1
`HIGH SPEED DATA TRANSMISSION USING
`EXPANDED BIT DURATIONS IN MULTIPLE
`PARALLEL CODED DATA STREAMS
`
`FIELD OF THE INVENTION
`
`This invention relates to a method for the high speed
`transmission of data, in particular in Wireless personal
`communications.
`
`BACKGROUND OF THE INVENTION
`
`During the last decade, the telecommunication industry
`has produced an explosion of Wireless technology. This
`growth, together With recent developments in hardWare
`miniaturiZation, has opened a neW dimension to future
`Wireless netWorks Whose ultimate goal is to provide univer
`sal personal communications. To achieve such an objective,
`the next generation personal communications netWorks Will
`need to be able to support a very high level of user traf?c
`along With a Wide range of high-quality services With
`varying bit rates. These future services are likely to include
`video and LAN applications Which require high speed
`transmission rates of several Mbps. HoWever, the ability to
`achieve high bit rates at loW error rates over Wireless
`channels is severely restricted by the propagation charac
`teristics of the Wireless environment Where signals typically
`arrive at the receiver via a scattering mechanism resulting in
`multiple propagation paths With different time delays,
`attenuation, and phasors. This causes a spread in delay times
`Which imposes a limit on the maximum transmission rate.
`These restrictions manifest themselves as intersymbol inter
`ference (ISI) Which leads to the introduction of an irreduc
`ible error ?oor. Therefore, Without countermeasures to miti
`gate the delay spread impairments the information rate is
`usually limited to be under 1 Mbps When user mobility
`prevents steady line-of-sight conditions.
`
`PRIOR ART
`
`One possible solution, Which has received a lot of atten
`tion recently, is multicarrier modulation or multitone modu
`lation in Which the transmitted data is divided into several
`interleaved bit streams Which are then used to modulate
`several sub-carriers, see for example:
`L. J. Cimini, Jr., “Analysis and simulation of a digital
`mobile-channel using orthogonal frequency division
`multiplexing,” IEEE Trans. Commun., COM-33, No. 7, pp.
`665—675, July 1985; I. Kalet, “The multitone channel,”
`IEEE Trans. Commun., Vol. COM-37, No.2, pp. 119—124,
`February 1989; and J. A. C. Bingham, “Multicarrier modu
`lation for data transmission: An idea Whose time has come,”
`IEEE Commun. Mag., pp. 5—14, May 1990.
`HoWever, such an approach requires equaliZation in the
`frequency domain Which can prove to be quite complex With
`high transmission rates (of the order of 10 Mbps or higher
`for instance) and time-varying Wireless channels. In
`addition, in multicarrier modulation a training Waveform is
`often sent through the channel. The channel information is
`then fed back from the receiver to the transmitter so that the
`total transmitted poWer for the various sub-channels can be
`allocated. To do so, hoWever, can be quite complicated. In
`fact, the optimum poWer distribution for each sub-channel
`should be calculated by a “Water-pouring” information
`theoretic approach similar to that of R. G. Gallagher “Infor
`mation Theory and Reliable Communications” Wiley, NY.
`1968. This again requires a complex system Whose perfor
`mance degrades signi?cantly if the channel feedback infor
`
`10
`
`15
`
`20
`
`25
`
`30
`
`35
`
`40
`
`45
`
`55
`
`60
`
`65
`
`5,960,032
`
`2
`mation is in error. The time-varying nature of the channel of
`Wireless channels further requires constant update of this
`complicated process. An alternative approach is to send
`sub-carrier pilots along With the transmitted information to
`assist phase and amplitude equaliZation in multicarrier
`modulation. HoWever, it should be noted that this reduces
`the spectrum efficiency.
`Another key advantage of multicode modulation is its
`ability to use interference cancellation as an effective tech
`nique for improving the overall system performance. Indeed,
`because of the special structure of the multicode modulation
`method there is only one channel for all data streams.
`Thereby, it is only necessary to estimate the channel param
`eters once Which can be done for instance by allocating a
`pilot channel for that purpose. (Note that if pilot signals are
`to be used in multicarrier modulation, then multiple pilot
`signals along With multiple channel estimations Will be
`required, thereby, making the system architecture quite
`complex.) In addition, in the multicode modulation method
`the signature sequences of all the data streams are knoWn.
`Hence, it folloWs that one can use the pilot signal to estimate
`the channel parameters. These parameters can then be used
`as an effective means for cancelling the interference among
`the sub-channels. Thereby, signi?cantly improving the over
`all system performance.
`
`SUMMARY OF THE INVENTION
`
`According to the present invention there is provided a
`method for the high speed transmission of data in a Wireless
`communication system comprising, dividing a high-rate data
`stream into a plurality of parallel loW-rate bit streams,
`Wherein each said loW-rate bit stream is modulated using
`direct-sequence spread spectrum as a single carrier.
`The choice of signature sequences or codes is crucial
`because these codes must be able to separate the interference
`betWeen the loW-rate bit streams and their multipath dupli
`cates. If both objectives in the code design can be achieved
`then the system sensitivity may be reduced to delay spread
`because of the spreading of the signaling interval and also
`take advantage of the inherent multipath rejection bene?t
`Without spreading the original bandWidth of the transmitted
`signal. As a result, the proposed method has the advantage
`of being more robust to fading and multipath problems than
`multicarrier modulation.
`Traditionally, PN sequences such as m-sequences,
`Kasami sequences, or Gold codes have been used in spread
`spectrum multiple access communications to separate the
`multiple users. HoWever, in the method of the present
`invention the loW-rate data streams are transmitted synchro
`nously. As a result orthogonal sequences, such as Walsh
`codes, Which have Zero cross-correlations When they are
`time synchronized, may be employed. HoWever, multipath
`delays can introduce signi?cant non-Zero cross-correlations
`betWeen orthogonal codes and therefore an alternative
`choice of signature sequence may be more appropriate.
`A preferred signature sequence may comprise a combi
`nation of orthogonal codes and PN sequences. Aparticularly
`preferred possibility is for the signature sequence of each
`loW-rate bit streams to be multiplied by the same PN code
`and then separated by different orthogonal sequence. By
`means of this arrangement the randomness of the orthogonal
`codes is increased While at the same time their Zero cross
`correlation property is maintained at Zero time delay.
`Preferably each said loW-rate bit streams is subject to a
`processing gain of the order of the number of said loW-rate
`bit streams. This makes it possible to obtain high-rate DS
`
`Page 9 of 15
`
`

`
`3
`spread spectrum modulation Within the bandwidth of the
`original high-rate transmission stream While maintaining the
`advantages of DS spread spectrum such as multi-path rejec
`tion.
`
`BRIEF DESCRIPTION OF THE DRAWINGS
`
`An embodiment of the invention Will noW be described by
`Way of example and With reference to the accompanying
`draWings, in Which:
`FIG. 1 illustrates schematically the method of data trans
`mission according to an embodiment of the present
`invention,
`FIG. 2 shoWs the BER performance of the present inven
`tion as a function of the normaliZed delay spread,
`FIG. 3 shoWs the BER performance of the present inven
`tion as a function of the spread spectrum resolution,
`FIGS. 4 and 5 shoW the BER performance of the present
`invention as a function of the normaliZed delay spread and
`for different numbers of loW-rate bit streams,
`FIG. 6 shoWs the effect on BER of selection diversity, and
`FIG. 7 illustrates a proposed interference cancellation
`method.
`
`DETAILED DESCRIPTION OF PREFERRED
`EMBODIMENT SYSTEM DESCRIPTION
`
`The proposed multicode modulation system is shoWn in
`FIG. 1. The incoming data bits With bit duration Tb are
`serial-to-parallel converted into K parallel bit streams With
`symbol duration T=KTb in a similar fashion to multitone
`modulation. After the serial-to-parallel conversion, the sym
`bols on each loW-rate branch are modulated using DS
`spread-spectrum modulation in Which the processing gain
`for each loW-rate stream is of the order of K. Consequently,
`it is possible to achieve high-rate DS spread spectrum
`modulation Within the bandWidth of the original high-rate
`transmission stream While maintaining the advantages of DS
`spread-spectrum such as multipath rejection. Furthermore,
`each of the DS spread-spectrum modulated loW-rate streams
`passes through exactly the same Wireless channel. As a
`result, poWer control Will not be an issue as it is for multiple
`access spread-spectrum systems. This common Wireless
`channel also implies that the received delay characteristics
`Will be identitical for all loW-rate data streams. Therefore,
`this makes the receiver design less intricate. For example, if
`a Rake receiver is required then the delay path search
`circuitry need only be implemented once rather than repeat
`edly for each individual loW-rate stream.
`In order to analyZe the performance of the multicode
`modulation technique shoWn in FIG. 1 some de?nitions and
`notation need to be introduced. Speci?cally, let Sk(t) denote
`the transmitted signal for the kth data stream. Then
`
`5m) = vmwmmw + so)
`
`(1)
`
`5,960,032
`
`4
`is the binary data signal, and
`
`w
`am) = Z agmm. (141m)
`
`(3)
`
`is the signature sequence signal With pt(t1, t2) being a unit
`rectangular pulse on [t1, t2), b]-(k)E{—1,1} Where Pr(b]-(k)=—
`1)=Pr(b]-(k)=1)=1/2, and the kth stream’s code sequence
`a]-(k)E{—1,1} With a]-(k)=a]-+N(k) for all j and k and for some
`integer N. The integer N is the minimum period of the
`spreading sequence. The chip length TC Will be assumed to
`be given by TC=T/N Where T is the symbol interval duration.
`Hence, there is one signature sequence ak=(a0(k), a1(k), .
`.
`. ,
`aN_1(k)) per data symbol.
`A multipath Rayleigh-fading channel having a sloW fad
`ing rate compared to the symbol rate may be assumed, so
`that the channel random parameters do not change signi?
`cantly over several consecutive symbol intervals. It may also
`be assumed that the channel consists of a ?xed number of
`faded paths. Speci?cally, the complex loWpass equivalent
`impulse response of the channel is given by
`
`Where L is the number of paths (A good approximation for
`L is L=[Am/TC]+1 Where Am is the maximum delay
`difference), and [31, "El, and Y1 are the lth path gain, delay, and
`phase for the kth data stream, and i=\/——1. Throughout, it may
`be assumed that for each 1 that the path phase of the received
`signal (bl, given by (methyl), to be an independent random
`variable uniformly, distributed over [0,275]. The path gains [31
`Will be assumed to be independent Rayleigh random vari
`ables.
`Auseful function Which characteriZes a multipath channel
`is the “poWer delay pro?le”
`
`pm % BMW].
`
`(5)
`
`15
`
`25
`
`35
`
`This function is important since it can provide a key param
`eter; namely, the (rms) delay spread A Which is de?ned as the
`square root of the second central moment. That is,
`
`45
`
`fpmdr
`
`(6)
`
`Where the average delay pd is given by
`
`55
`
`frpmd r
`fpmdr
`
`-
`
`M =
`
`d
`
`Where P is the signal power, we is the carrier frequency, and
`11) is the carrier phase. Likewise,
`
`It turns out that the bit error rates (BERs) for transmission
`through a multipath channel are strongly dependent on the
`normaliZed rms delay spread Which is de?ned as
`
`65
`
`Where We recall that T is the symbol period. In fact, d must
`be signi?cantly smaller than unity in order to prevent
`excessive irreducible BERs due to ISI in a conventional
`(K=1) non-equaliZed modulation scheme.
`
`Page 10 of 15
`
`

`
`5,960,032
`
`5
`Throughout, We Will restrict our consideration to the
`equal-gain tWo-ray pro?le
`
`1
`PU) = 5W1) + 6(1 - 2A)].
`
`<8)
`
`6
`-continued
`
`Rlk’w = fram- 011mm and
`
`0
`T
`
`Rlk’w = f aka-mm:
`
`(12)
`
`In this case, We have L=2, pd=A and dEA/T. In addition, note
`that at a speci?c time delay t, the channel impulse response,
`denoted by h(t,), is a complex Zero-mean Gaussian random
`variable With variance p(t,).
`NoW by combining Equations (1) and (4) and using the
`convolutional integral, the received signal, Which Will be
`denoted as r(t), can then be expressed as
`
`15
`
`, K. It can be easily shoWn that
`.
`.
`for 0§T<T and k=1, 2, .
`the continuous-time partial correlation functions de?ned in
`(12) can be expressed as
`
`Where l=|_'C/TCJ and C1(k)(l) is the aperiodic cross-correlation
`function. That is,
`
`(13)
`
`Where n(t) is the channel noise Which is assumed to be a
`White Gaussian noise process With double sided poWer
`spectral density No/2.
`
`6%) =
`
`ERROR PERFORMANCE ANALYSIS
`
`Without loss of generality, We shall restrict our consider
`ation to the 1st loW-rate data stream. Further, We Will assume
`that the desired receiver can coherently recover the carrier
`phase and delay lock to the jth path of the arriving desired
`signal. Therefore, if We let Zj- denote the output of the
`correlator receiver that is matched to the jth path of data
`stream 1. (When there are more than 2 signi?cant paths, a
`Rake receiver structure With multiple correlators must be
`considered.) Then, the decision statistic is given by
`(assuming that the maximum delay difference is less than
`tWo symbols)
`
`25
`
`35
`
`45
`
`55
`
`and C1(k)(l)=0 for |l|§N. Also note that since n(t) is a White
`noise Gaussian process, it folloWs that 11 is a Zero mean
`Gaussiani random variable With variance NOT/4.
`NoW let Pg denote the system BER for the ?rst loW-rate
`data stream. Then
`
`Due to the symmetry of the problem, one can Without loss
`of generality set b0(1)=—1. Hence, PeEP(Z§0|b0(1)=—1).
`Speci?cally, (for simplicity, We Will omit the conditioning
`on boa) in our notation)
`
`Where
`
`for l=1, 2,. .
`
`. , L and k=1, 2,. .
`
`.
`
`, K. Next let
`
`A
`L
`S: 2 x51)
`l:lJ¢j
`
`denote the desired user “self” interference, and
`
`A K L
`
`k
`
`22x1)
`
`I<:2 [:1
`
`(16)
`
`(17)
`
`Where tl='cl—"cj- and m=0 or 1 depending on Whether tIZO or
`tl<0. Likewise,
`
`denote the total co-channel interference. Likewise, let the
`random vector ®=(B,D,(I>,b) Where B=([31,[32, .
`.
`. ,BK),
`D=("c1;c2, .
`.
`. ,"cK), <I>=(q>,,q>,, .
`.
`. ,¢K), and b=(b1,b3, .
`.
`. ,bK)
`Where bk=(bm_1(k),bm(k)) for k=1,2, .
`.
`. ,K and m=0 or 1
`again depending on Whether the delay difference, t], is
`positive or negative as described above. Then, for a given 6)
`
`65
`
`Page 11 of 15
`
`

`
`5,960,032
`
`7
`(With b0(1)E—1) the decision statistic Z]- is conditionally
`Gaussian. In particular,
`
`Where 113(6) is the joint probability “density” function of 6)
`(Note that the term “density” is being used in a loose
`statistical sense as the random vector 6 contains discrete and
`continuous random variables.) and
`
`PE(0) = Pr(error| o = 0)
`
`10
`
`(19)
`
`15
`
`25
`
`35
`
`Where Eb=PT is the energy per bit. Notice that if [3,=1 for all
`I and K=L=1, then We have no self nor co-channel interfer
`ence and Eqn. (19) reduces to the Well-known BPSK result
`over the additive White Gaussian noise (AWGN) channel,
`
`a
`
`is the complementary error function.
`Where
`The probability of error as expressed in Equation (18)
`cannot be in practical situations evaluated analytically. As a
`result, Monte Carlo simulations must be often used to
`estimate Pg. Let Ie(Z]-) denote the indicator random variable
`of the error event {Z120}. That is, Ie(Z]-)=1 if Z120.
`QtherWise, Ie(Z]-)=0. Then the Monte Carlo estimator for P6,
`P6, is the sample mean estimator Which simply counts the
`relative frequency of the event {Z120} during N indepen
`dent simulation trials and then estimates Pg as folloWs
`
`1 N
`e = 52142;”)
`
`(20)
`
`, Zj-(N) are N independent and
`.
`.
`Where Zj-(l), ZJ-(Z), .
`identically distributed (iid) random samples that are gener
`ated during the simulation trials. The Monte Carlo estimator
`as described in Equation (20) often requires a large number
`of simulation trials. This is especially true When the BERs
`are relatively small. Amore ef?cient approach for estimating
`Pg can be obtained if one uses Eqns. (18) and (19). Indeed,
`notice that
`
`45
`
`N
`
`1
`i), E Z mow)
`
`2 |
`n:l
`
`(21)
`
`55
`
`. ., N
`Where Pe(®(”)) is given by (19) and GO’) for n=1,2, .
`are iid random samples from 366(6). Clearly, the computa
`tional cost required to achieve accurate estimates of P6 can
`be signi?cantly decreased through the use of Equation (21)
`instead of Equation (20).
`NoW it is clear that the overall system performance is
`dependent on the choice of the signature sequences.
`Traditionally, PN sequences such as m-sequences, Kasami
`sequences, or Gold codes have been used in spread
`spectrum multiple access communications to separate the
`multiple users. HoWever, in this application the loW-rate data
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`streams are transmitted synchronously. As a result, since
`orthogonal sequences such as Walsh codes have Zero cross
`correlations When they are time synchroniZed, one Would
`expect that orthogonal signature sequences can be applied in
`multicode modulation. But, because multipath delays can
`introduce signi?cant non-Zero cross-correlations betWeen
`the orthogonal codes another choice of signature sequences
`may be more appropriate. A possible alternative is to use a
`combination of orthogonal codes and PN sequences.
`Speci?cally, the symbol sequences for the K data streams
`can be multiplied by the same PN code and then separated
`by different orthogonal sequences. By doing so, the random
`ness for the orthogonal codes is increased While at the same
`time their Zero cross-correlation property is maintained at
`Zero time delay. Thus, the signature sequence aj-(k) can be for
`example, an orthogonal sequence (eg Walsh codes), or an
`orthogonal sequence multiplied or “concatenated” by a PN
`code Which is the same for all data streams.
`
`NUMERICAL RESULTS
`In this section, some sample numerical results are pre
`sented Which illustrate the potential of the proposed multi
`code modulation method of the present invention as applied
`to the transmission of high bit rates in Wireless personal
`communications. Throughout, We Will use as an example a
`bit rate Rb=10 Mbps. Hence, the loW information rate of the
`parallel data streams is simply R=10/K Mbps. Recall that
`our results are speci?c to the L=2 delay pro?le given by
`Equation
`HoWever, We note that it has been shoWn that
`the performance of various communications systems over
`multipath fading channels is not very sensitive to the delay
`pro?le used. In fact, if L22 and if a Rake receiver With
`maximum ratio combining among the various signals from
`multiple correlators is considered, better results may be
`obtained at the expense of higher complexity. Finally, We
`notice that in this study We have restricted our consideration
`to the high signal-to-noise (SNR) or the irreducible BER
`performance. In other Words, our main interest is the “irre
`ducible” bit errors Which occur at very high SNRs. These
`errors typically occur because of signal fading and/or ISI
`caused by multipath delay spread.
`FIG. 2 lists the BER performance of the proposed mul
`ticode modulation scheme as a function of the normaliZed
`delay spread d When Eb/N0=40 dB. Speci?cally, this ?gure
`lists the average bit error probability using Walsh sequences
`With K=N=64, m-sequences With period N=63 along With 63
`loW-rate parallel data streams, and K=64 Walsh sequences
`that are multiplied or concatenated With the ?rst 64 bits of
`an m-sequence With length 127. A close observation of this
`?gure clearly indicates the potential of the method of the
`present invention for the transmission of high-speed data.
`For example, note that When d=0.1 We have Rb=1/TC=0.1><
`108, R=0.1562><106, and A=0.64><10_6=640 nsec, Which is
`approximately the upper bound of the normaliZed rms delay
`spread in the personal communications environment. For
`comparison, it is noted that With this value of A the normal
`iZed rms delay spread is equal to 6.4 When K51 (i.e., over a
`single user channel). Obviously, the transmission over the
`channel With small error rates in this latter case is impossible
`unless one uses an alternative modulation scheme such as
`the one proposed here.
`The importance of selecting appropriate signature
`sequences is also clearly indicated in FIG. 2. Speci?cally,
`note that for large delay spreads the PN sequences give the
`best performance. HoWever, When the delay spread is rela
`tively small multicode modulation With PN sequences is
`unusable. In contrast, note that the orthogonal Walsh
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`sequences give the best and Worst performance among the
`three considered codes for small values and large values of
`d, respectively. Finally, note that the performance of the
`concatenated Walsh/m-sequences is a compromise of the
`above coding schemes. This can be easily explained by
`presenting FIG. 3 Which shoWs the BERs listed in FIG. 2 as
`a function of the DS spread-spectrum modulation “resolu
`tion” Which We Will de?ne as
`
`Clearly, the performance of the PN sequences is expected to
`be very bad for values of ré 1 since in this case the delayed
`paths cannot be resolved. For these values, the time delays
`are relatively small. As result, the orthogonal Walsh func
`tions Will give the best performance because of their small
`cross-correlations Which makes the co-channel interference
`nearly Zero. In contrast, for large values of d the non-Zero
`cross-correlations of Walsh sequences are not negligible.
`Consequently, the performance of multicode modulation
`With Walsh codes is Worse than that With PN sequences.
`Finally, note that the concatenated Walsh/PN-sequences
`combine the good features of orthogonal codes as Well as PN
`sequences. Speci?cally, they solve the cross-correlation
`problem of the Walsh codes When the delay spread is large
`and maintain the “orthogonality” property of the Walsh
`sequences When the delay spread is small.
`NoW note that the performance of the multicode modu
`lation method is not as good as one Would hope for. This is
`expected because the performance of the system over a
`single Rayleigh faded path (i.e., With K=L=1) is poor to
`begin With. Indeed, in this case our simulations predict that
`P€=0.23><10_5. By using a fully loaded spread spectrum
`system (i.e., a DS spread spectrum Where the total number
`of simultaneous users is equal to K) one Would expect that
`the performance Would degrade and becomes relatively
`poor. One possible solution is to use less loW-rate parallel
`data streams as shoWn in FIG. 4. This Will clearly improve
`the system performance (as there Will be less co-channel
`interference) at the expense of the channel bandWidth usage.
`This is also illustrated in FIG. 5 Which shoWs the average
`error probability against d When Walsh codes With N=128
`are concatenated With the ?rst 128 chips of an m-sequence
`With period 255.
`The performance of spread spectrum multiple-access over
`multipath fading channels can be signi?cantly improved if
`one employs some type of diversity such as Selection
`diversity or Maximal Ratio Combining diversity. This is
`illustrated in FIG. 6 Which lists the BER performance When
`antenna selection diversity of order M=2 is used. It Was also
`found that performance among sub-channels is different, ie
`“code-selective” fading exists, similar to frequency
`selective fading that occurs in the multicarrier modulation.
`It is therefore expected that coding and interleaving could
`help reduce the associated bursty errors.
`The results presented here indicate that multicode modu
`lation can indeed be a potential candidate for high-speed
`transmission. To further improve the system performance
`and because of the special structure of multicode modulation
`one can cancel the interference among the sub-channels
`using an interference cancellation method. The proposed
`interference cancellation method is based on a successive
`co-channel interference cancellation scheme Whose aim is to
`improve the system performance by incorporating the avail
`able information about the interference signals in the deci
`sion process. This is done by regenerating estimates of the
`interfering signals, and then subtracting those reconstructed
`interference signals from the input of the desired receiver.
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`This process is performed in a cascaded fashion in such a
`Way that more and more of the interference signals are
`cancelled, thereby resulting in a signi?cant performance
`improvement. This process is performed successively (i.e.,
`in multiple stages) as folloWs (See FIG. 7).
`We ?rst ?nd initial estimates of the co-channel interfer
`ence terms (stage O). We Will then proceed With m stages of
`processing the decision statistics of the desired signal. At the
`1 stage, We reconstruct the co-channel interference terms
`In[l] Where n denotes the index of the nth bit interval using
`estimates of the channel (namely, the multipath time delay
`"Ac, please delay (I), and path gain
`along With estimates of
`the unknoWn transmitted symbols of data stream k, bn(k) and
`then subtract the regenerated co-channel interference for the
`desired data stream received signal to obtain a neW decision
`statistic for the lth stage. Ideally, if We can correctly estimate
`the co-channel interference terms at say stage I, then the
`decision statistic at the next stage Will not include any
`co-channel interference terms. Consequently, signi?cant
`improvement in the system performance Would result. On
`the other hand, note that if the data estimation for say data
`stream k is incorrect at say stage I, then a negative version
`of the interference caused by this data stream is created,
`thereby, resulting in a neW co-channel interference term (due
`to data stream k) at the 1+1 stage that is tWice the co-channel
`interference term of data stream k at stage I. It is therefore
`clear that one must have relatively good data and channel
`estimates in order for this iterative scheme to Work Well.
`HoWever, note that because of the special structure of the
`multicode modulation system, there is only one channel for
`all the data streams. Hence, We only need to estimate the
`channel parameters once. This can be done for instance by
`allocating a pilot channel for that purpose. Further, notice
`that the signature sequences, ak(t), of all the data streams are
`knoWn. Hence, it is reasonable to expect that the proposed
`interference cancellation method to perform Well in the
`multicode modulation system. Such performance improve
`ment has been actually observed and Was found to be true
`during various simulations that have been conducted.
`We claim:
`1. A method for transmitting digital data in a Wireless
`communication environment comprising:
`dividing an incoming stream of serial data bits having a
`?rst bit duration (Tb) into a plurality
`of parallel data
`bit streams;
`expanding by K times the bit duration of the incoming
`data so that the resulting symbol duration in said
`parallel data streams equals KTb;
`modulating said expanded parallel data streams With
`modulating sequences, each said modulating sequence
`having a processing gain N, having a sequence period
`equal to the symbol duration KTb of said expanded data
`streams, and having N binary chips Within each period
`so that each chip has a chip duration of TC=KTb/N,
`Wherein K and N are integers and N>K; and
`summing the modulated parallel data streams for trans
`mission.
`2. A method according to claim 1, Wherein said modu
`lating sequences are mutually orthogonal.
`3. A method according to claim 1, Wherein said modu
`lating sequences are pseudo-noise (PN) sequences.
`4. The method of claim 3, Wherein said PN sequences are
`selected from the group comprising m-sequences, Kasami
`sequences and Gold sequences.
`5. A method according to claim 1, Wherein each said
`modulating sequence is a combination of a pseudo-noise
`sequence and an orthogonal sequence.
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`11
`6. A method according to claim 5 wherein said combina
`tion is formed by multiplication or concate