TNY264I266-268
`
`Tinyswitchm-ll Family
`Enhanced, Energy Efficient,
`Low Power Off-line Switcher
`
`Product Highlights
`
`Tinyswitch-ll Features Reduce System Cost
`- Fully integrated auto-restart for short circuit and open
`loop fault protection - saves external component costs
`- Built-in circuitry practically eliminates audible noise with
`standard varnished transformer
`
`- Programmable line under—voltage detect feature prevents
`power on/off glitches - saves external components
`- Frequency jittering dramatically reduces EMI (~10 dB)
`- minimizes EMI filter component costs
`132 kHz operation reduces transformer size - allows use
`of EFl2.6 or EE13 cores for low cost and small size
`
`- Very tight tolerances and negligible temperature variation
`on key parameters eases design and lowers cost
`- Lowest component count switcher solution
`
`Better CostIPerformance over RCC & Linears
`
`- Lower system cost than RCC, discrete PWM and other
`integrated/hybrid solutions
`- Cost effective replacement for bulky regulated linears
`- Simple ON/OFF control — no loop compensation needed
`- No bias winding — simpler, lower cost transformer
`
`Extremely Energy Efficient
`~ No load consumption < 50 mW with bias winding and
`< 250 mW without bias winding at 265 VAC input
`- Meets Blue Angel, Energy Star, and EC requirements for
`standby power consumption
`ideal for cell—phone charger and PC standby applications
`
`High Performance at Low Cost
`- High voltage powered — ideal for charger applications
`- High bandwidth provides fast turn on with no overshoot
`- Current
`limit operation rejects line frequency ripple
`- Built-in current limit and thermal protection
`
`Description
`
`TinySwitch-[1 maintains the simplicity of the TinySwitch
`topology, while providing a number of new enhancements to
`further reduce system cost, component count and audible noise.
`Like Tz'rzySwitch, a 700 V power MOSFET, oscillator, high
`voltage switched current source, current limitandthermal shutdown
`circuitry are integrated onto a monolithic device. The start-up
`and operating power are derived directly from the voltage on the
`DRAIN, eliminating the need for a transformer bias
`
`INTEGRATIONS INC.
`Pl-2684401700
`
`
`PRODUCT“
`
`TNY264P or G
`
`TNY266P or G
`
`TNY267P or G
`TNY268P or G
`
`6
`
`10W
`
`Table 1. Notes: 1. Typical continuous power in a non-ventilated
`enclosed adapter measured at 50 °C ambient. 2. Maximum
`practical continuous power in an open frame design with adequate
`heat sinking, measured at 50 °C ambient (See key applications
`section for details). 3. Packages: P: DIP-8B, G: SMD-SB. Please
`see part ordering information.
`
`TinySwitc/1-II devices incorporate auto-restart, line under-
`voltage sense, and frequency jittering. An innovative design
`minimizes audio frequency components in the simple ON/OFF
`control scheme to practically eliminate audible noise with
`standard taped/varnished transformer construction. The fully
`integrated auto—restar1 circuit safely limits output power during
`fault conditions such as output short or open loop, reducing
`component count and secondary feedback circuitry cost. The
`line under-voltage sense threshold can be externally programmed
`. using an optional line sense resistor, eliminating power down
`glitches caused by the slow discharge ofinput storage capacitors
`present in applications such as standby supplies. The operating
`frequency of 132 kHz is j ittcred to significantly reduce both the
`
`POWER
`
`‘
`
`-
`
`—
`
` +
`
`WAC 1006
`
`

`
`REGULATOR
`
`VB
`
`YPASS PIN
`
`LINE U NDER-VOLTAGE
`
`AUTO-
`RESTART
`COUNTER
`
`CURRENT
`LIMIT STATE
`MACHINE
`
`TNY264I266-268
`
`avmss
`(BP)
`0
`
`Z40).LA l 3
`
`ENABLE]
`UNDER-
`VOLTAGE
`(ENIUV)
`
`LEADING
`EDGE
`BLANKING
`
`SOURCE
`(S)
`Pl-2643-091800
`
`Figure 2. Functional Block Diagram.
`
`Pin Functional Description
`
`.
`DRAIN (D) Pin:
`Power MOSFET drain connection. Provides internal operating
`current for both start-up and steady-state operation.
`
`BYPASS (BP) Pin:
`Connection point for a 0.1 pF external bypass capacitor for the
`internally generated 5.8 V supply.
`
`ENABLE/UNDER—VOLTACE (EN/UV) Pin:
`This pin has dual functions, enable input and line under—Voltage
`sense. During normal operation, switching of the power
`MOSFET is controlled by this pin. MOSFET switching is
`terminated when a current greater than 240 LLA is drawn out of
`this pin under most loads. However, at high load levels, even
`when more than 240 pm is drawn out of this pin the MOSFET
`switching still occurs, but at a predetermined lower current limit
`level. This pin also senses line under-voltage conditions through
`an external resistor connected to the DC line voltage. If there
`is no external resistor connected to this pin, TinySwitch~1I
`detects this and disables the line under-voltage function.
`
`P Package (DIP-8B)
`G Package (SMD-8B)
`
`ENIUV fl Pl-2685-101600
`
`s (HV RTN)
`
`S (HV RTN)
`
`Figure 3. Pin Configuration.
`
`SOURCE (S) Pin:
`Control circuit common,
`MOSFIET source.
`
`SOURCE (HV RTN) Pin:
`
`internally connected to output
`
`

`
`
`
`TNY264l266-268
`
`Tinyswitch-ll Functional Description
`
`TinySwitch-II combines a high voltage power MOSFET switch
`with apower supply controller in one device. Unlike conventional
`PWM (Pulse Width Modulator) controllers, TinySwitch-II uses
`a simple ON/OFF control to regulate the output voltage.
`
`The TinySwiz‘ch-11 controller consists of an Oscillator, Enable
`Circuit (Sense and Logic), Current Limit State Machine, 5.8 V
`Regulator, Bypass pin Under-Voltage Circuit, Over
`Temperature Protection, Current Limit Circuit, Leading Edge
`Blanking and a 700 V power MOSFET. TinySwitc}2—II
`incorporates additional circuitry for Line Under-Voltage Sense,
`Auto-Restart andFrequency Jitter. Figure 2 shows the functional
`block diagram with the most important features.
`
`Oscillator
`
`The typical oscillator frequency is internally set to an average
`of 132 kHz. Two signals are generated from the oscillator, the
`Maximum Duty Cycle signal (DCMAX) and the Clock signal that
`indicates the beginning of each cycle.
`
`The Tz'nySwitch-II oscillator incorporates circuitry that
`introduces a small amount of frequency jitter, typically 8 kHz
`peak—to-peak, to minimize EMI emission. The modulation rate
`ofthe frequencyjitter is set to 1 kHz to optimize EMI reduction
`for both average and quasi-peak emissions. The frequency jitter
`should be measured with the oscilloscope triggered at the
`fall ing edge ofthe DRAIN waveform. The waveform in Figure 4
`illustrates the frequency jitter of the TinySwz’tch-II.
`
`Enable Input and Current Limit State Machine
`The enable input circuit at the EN/UV pin consists of a low
`impedance source follower output set at 1.0 V. The current
`through the source follower is limited to 240 uA. When the
`current drawn out ofthis pin exceeds 240 uA, a low logic level
`
`100
`
`5
`
`Time (us)
`
`10
`
`Figure 4. Frequency Jitter.
`
`(disable) is generated at the output of the enable circuit. This
`enable circuit output is sampled at the beginning of each cycle
`on the rising edge oftheclock signal. Unclermost load conditions,
`ifhigh, the power MOSFET is turned on for that cycle (enabled),
`otherwise the power MOSF ET remains off (disabled). At very
`high loads, even when a low logic level is generated at the
`output ofthe enable circuitry, the power MOSFETis still turned
`on for that cycle (enabled), but the current limit is reduced to a
`predetermined level (typically 50% of the specified current
`limit). Since the sampling is done only at the beginning ofeach
`cycle, subsequent changes in the EN/UV pin voltage or current
`during the remainder of the cycle are ignored.
`
`The Current Limit State Machine reduces the current limit by
`discrete amounts at light loads when TI'nySwitch-II is likely to
`switch in the audible frequency range. The lower current limit
`reduces the transformer flux density and the associated audible
`noise. At very high loads, the state machine prevents cycle
`skipping and thus reduces the audible frequency components
`associated with it. The state machine monitors the sequence of
`EN/UV pin voltage levels to determine the load condition and
`adjusts the current limit level accordingly in discrete amounts.
`
`Under most operating conditions (except when close to no-
`load), the low impedance of the source follower keeps the
`voltage on the EN/UV pin from going much below 1.0 V in the
`disabled state. This improves the response time ofthe optocoupler
`that is usually connected to this pin.
`
`5.8 V Regulator and 6.3 V Shunt Voltage Clamp
`The 5.8 V regulator charges the bypass capacitor connected to
`the BYPASS pin to 5.8 V by drawing a current from the voltage
`on the DRAIN, whenever the MOSFET is off. The BYPASS
`pin is the internal supply voltage node for the TinySwitc/1-11.
`When the MOSFET is on, the TinySwilch—1I runs off of the
`energy stored in the bypass capacitor. Extremely low power
`consumption of the internal circuitry allows TinySwitch-II to
`operate continuously from the current drawn from the DRAIN
`pin. A bypass capacitor value of 0.1 ul? is sufficient for both
`high frequency decoupling and energy storage.
`
`ln addition, there is a 6.3 V shunt regulator clamping the
`BYPASS pin at 6.3 V when current is provided to the BYPASS
`pin through an external resistor. This facilitates powering of
`TinySwizc/1-11 externally through a bias winding to decrease the
`no load consumption to about 50 mW.
`
`BYPASS Pin Under—Voltage
`The BYPASS pin under-voltage circuitry disables the power
`MOSFET when the BYPASS pin voltage drops below 4.8 V.
`Once the BYPASS pin voltage drops below 4.8 V, it must rise
`back to 5.8 V to enable (turn-on) the power MOSFET.
`
`

`
`TNY264I266-268
`
`Over Temperature Protection
`The thermal shutdown circuitry senses the die temperature. The
`threshold is set at 135 “C with 70 °C hysteresis. When the die
`temperature rises above this threshold (135 °C) the power
`MOSFET is disabled and remains disabled until
`the die
`
`temperature falls by 70 °C, at which point it is re-enabled.
`
`Current Limit
`
`The current limit circuit senses the current in the powerMOSFET.
`When this current exceeds the internal threshold (ILMT), the
`power MOSFET is turned off for the remainder of that cycle.
`The current limit state machine reduces the current limit threshold
`
`by discrete amounts under medium and light loads.
`
`The leading edge blanking circuit inhibits the current limit
`comparator for a short time (tLEB) after the power MOSFET is
`turned on. This leading edge blanking time has been set so that
`current spikes caused by capacitance and secondary-side rectifier
`reverse recovery time will not cause premature termination of
`the switching pulse.
`
`Auto-Restart
`
`In the event of a fault condition such as output overload, output
`short, or an open loop condition, Tz'nySwirch-ll enters into auto-
`restart operation. An internal counter clocked by the oscillator
`gets reset every time the EN/UV pin is pulled low. if the
`EN/UV pin is not pulled low for 50 ms, the power MOSFET
`switching is disabled for 850 ms (except in the case of line
`under-voltage condition in which case it is disabled until the
`condition is removed). The auto—restart alternately enables and
`disables the switching of the power MOSFET until the fault
`condition is removed. Figure 5 illustrates auto-restart circuit
`operation in the presence of an output short.
`
`In the event of a line under-voltage condition, the switching of
`the power MOSFET is disabled beyond its normal 850 ms time
`until the line under-voltage condition ends.
`
`Pl-2099-01
`
`2401
`
`Line Under—Voltage Sense Circuit
`The DC line voltage can be monitored by connecting an
`external resistor from the DC line to the EN/UV pin. During
`power-up or when the switching of the power MOSFET is
`disabled in auto-restart, the current into the EN/UV pin must
`exceed 50 uA to initiate switching of the power MOSFET.
`During power-up, this is implemented by holding the BYPASS
`pin to 4.8 V while the line under—voltage condition exists. The
`BYPASS pin then rises from 4.8 V to 5.8 V when the line under-
`voltage condition goes away. When the switching ofthe power
`MOSFET is disabled in auto—restart mode and a line under-
`
`voltage condition exists, the auto—restart counter is stopped.
`This stretches the disable time beyond its normal 850 ms until
`the line under—voltage condition ends.
`
`The line under-voltage circuit also detects when there is no
`external resistor connected to the EN/UV pin. In this case the
`line under—voltage function is disabled.
`
`Tinyswitch-ll Operation
`
`TinySwitch—II devices operate in the current limit mode. When
`enabled, the oscillator turns the power MOSFET on at the
`beginning of each cycle. The MOSFET is turned off when the
`current ramps up to the current limit or the DCMAX limit is
`reached. As the highest current limit level and frequency of a
`TinyS'witch~/'1 design are constant, the power delivered to the
`load is proportional to the primary inductance ofthe transformer
`and peak primary current squared. Hence designing the supply
`involves calculating the primary inductance of the transformer
`for the maximum output power required. Ifthe TinySwitc/1-11 is
`appropriately chosen for the power level, the current in the
`calculated inductance will ramp up to current limit before the
`
`DCMAX limit is reached.
`
`Enable Function
`
`TinySwitch-11 senses the EN/UV pin to determine whether or
`not to proceed with the next switch cycle as described earlier.
`The sequence of cycles is used to determine the current limit.
`Once a cycle is started, it always completes the cycle (even
`when the EN/U V pin changes state halfway through the cycle).
`This operation results in a power supply in which the output
`voltage ripple is determined by the output capacitor, amount of
`energy per switch cycle and the delay of the feedback.
`
`is generated on the secondary by ‘
`The EN/UV pin signal
`comparing the power supply output voltage with a reference
`voltage. The EN/UV pin signal is high when the power supply
`output voltage is less than the reference voltage.
`
`In a typical implementation, the EN/UV pin is driven by an
`optoeoupler. The collector of the optoeoupler transistor is
`connected to the EN/UV pin and the emitter is connected to the
`SOURCE pin. The optoeoupler LED is connected in series with
`a Zener across the DC output voltage to be regulated. When the
`
`

`
`TNY264l266-268
`
`the sequence of samples over multiple cycles, it determines the
`appropriate current limit. At high loads, when the EN/UV pin
`is high (less than 240 p.A pulled out ofthe pin), then a switching
`cycle with the full current limit occurs. At lighter loads, when
`EN/UV is high, then a switching cycle with a reduced current
`limit occurs.
`
`Under most load conditions, when the FN/U V pin is low (more
`than 240 }1A pulled out of the pin), no switching cycle occurs.
`However, at very high loads, when the EN/UV pin is low, a
`switching cycle with reduced current limit occurs. The EN/UV
`pin status is sampled again at the start ofthe subsequent clock
`cycle.
`
`At nearly full load, TinySwitc/1-II will conduct during all of its
`clock cycles (Figure 6). EN/UV only modulates the current
`limit between two distinct levels. At slightly lower loads, it will
`“skip” cycles in order to maintain voltage regulation at the
`power supply output (Figure 7). At medium loads, cycles will
`be skipped and the current limit will be reduced (Figure 8). At
`very light loads, the current limit will be reduced even further
`(Figure 9). Only a small percentage of cycles will occur to
`satisfy the power consumption of the power supply.
`
`The responseti/me ofthe TinySwitch—1ION/OFF control scheme
`is very fast compared to normal PWM control. This provides
`tight regulation and excellent transient response.
`
`Power Up/Down
`The TinySwitch-II requires only a 0.1 ttF capacitor on the
`BYPASS pin. Because of the small size of this capacitor, the
`time to charge this capacitor is kept to an absolute minimum,
`typically 0.6 ms. Due to the fast nature ofthe ON/OFF feedback,
`
`
`
`PI-2377-091100
`
`Figure 8. YinySwi1ch—11 Operation at Medium Load.
`
`output voltage exceeds the target regulation voltage level
`(optocoupler LED voltage drop plus Zener voltage), the
`optocoupler LED will start to conduct, pulling the EN/UV pin
`low. The Zener can be replaced by a TL43l reference for
`improved accuracy.
`
`ON/OFF Operation with Current Limit State Machine
`The internal clock ofthe TinySwitch-II runs all the time. At the
`beginning of each clock cycle, it samples the EN/UV pin to
`decide whether or not to implement a switch cycle and based on
`
`
`
`Pl-2667-090700
`
`Figure 6. 77nySwitch—I] Operation at Very Heavy Load.
`
`Pl«266D-072400
`
`

`
`TNY264l266—268
`
`200
`
`100
`
`200
`
`400
`
`Time (ms)
`
`Figure I 1. TinySwitch—lI Power-up Without External
`Resistor Connected to EN/UV Pin.
`
`
`
`Pl-234B~01059B
`
`Pl-2395-010599
`
`
`
`Pl-2661-4372400
`
`Figure 9. 7inySwitch-I1 Operation at Very Light Load.
`
`there is no overshoot at the power supply output. When an
`external resistor (2 M9) is connected to the EN/UV pin, the
`power MOSFET switching will be delayed during power-up until
`the DC line voltage exceeds the threshold ( 1 00 V). Figures 10 and
`11 show the power-up timing waveform of TinySwitcl1-II in
`applicationswith andwithoutan external resistor (2 M9) connected
`to the EN/UV pin.
`
`During power-down, when an external resistor is used, the
`power MOSFET will switch for 50 ms after the output loses
`regulation. The power MOSFET will then remain offwithout
`any glitches since the under-voltage function prohibits restart
`when the line voltage is low.
`
`Pl-2333-1
`
`22396
`
`Time (ms)
`Figure 10. TinySwitch~II P0wer—up With External Resistor
`
`Figure 13. Slow Power-down Timing with External (2 M12)
`
`Time (s)
`
`

`
`TNY264l266—268
`
`ca 680 pF
`Y1 Safety
`
`Optional line
`UV detect resistor
`
`1N5819
`
`TNY264
`
`Tinyswitch-ll
`
`Pl-2706-012901
`
`Figure 14. 2.5 W Constant Current, Constant Voltage Battery Charger with Universal Input (85 - 265 VAC).
`
`Figure 12 illustrates a typical power—down timing waveform of
`TinySwitch-I1. Figure 13 illustrates a very slow power-down
`timing waveform of TinySwitch-11 as in standby applications.
`The external resistor (2 M9) is connected to the EN/UV pin in
`this case to prevent unwanted restarts.
`
`The TinySwitch—[I does not require a bias winding to provide
`power to the chip, because it draws the power directly from the
`DRAIN pin (see Functional Description above). This has two
`main benefits. First, for a nominal application, this eliminates
`the cost of an extra bias winding and associated components.
`Secondly, for charger applications, the current-voltage
`characteristic often allows the output voltage to fall to low
`values while still delivering power. This type of application
`normally requires a forward-bias winding which has many
`more associated components. With Tiny-Switch—II, neither are
`necessary. For applications that require a very small power
`consumption (50 mW), a resistor from a bias winding to the
`BYPASS pin can provide the power to the chip. The BYPASS
`pin in this case will be clamped at 6.3 V. This method will
`eliminate the drawing of the power from the DRAIN pin,
`thereby reducing the no-load power consumption and improving
`full-load efficiency.
`
`Current Limit Operation
`Each switching cycle is terminated when the DRAIN current
`reaches the current limit of the TinySwitch~II. For a given
`primary inductance and inputvoltage, the duty cycle is constant.
`
`However, the duty cycle does change inversely with the input
`voltage providing “voltage feed—forward” advantages: good
`line ripple rejection and relatively constant power delivery
`independent of the input voltage.
`
`BYPASS Pin Capacitor
`The BYPASS pin uses a small 0.1 1.LF ceramic capacitor for
`decoupling the internal power supply of the TinySwitch—11.
`
`Application Examples
`
`The TinySwitch-II is ideal for low cost, high efficiency power
`supplies in a wide range of applications such as cellular phone
`chargers, TV standby, AC adapters, motor control, appliance
`control and ISDN network termination. The 132 kHz operation
`allows the use of a low cost EEI3 or F.Fl2.6 core transformer
`
`while still providing good efficiency. The frequency jitter in
`TinySwitch-[1 makes it possible to use a single inductor (or two
`small resistors if lower efficiency is acceptable) in conjunction
`with two input capacitors for input EMI filtering. The auto-
`restart function removes the need to oversize the output diode
`for short circuit conditions allowing the design to be optimized
`for low cost and maximum efficiency. In charger applications,
`it eliminates the need for a second optocoupler and Zener diode
`for open loop fault protection. Auto—restart also saves the cost
`of adding a fuse or increasing the power rating of the current
`sense resistors to survive reverse battery conditions. For
`
`

`
`TNY264I266-268
`
`TinySwz'tch-II eliminates several components and saves cost.
`TinySwitch—1Iis well suited for applications that require constant
`voltage and constant current output. As TinySwitch—II is
`always powered from the input high voltage, it therefore does
`not require a bias winding for operating power. Consequently,
`its operation is not dependant on the level ofthe outputvoltage.
`This greatly simplifies designing chargers that must work
`down to zero volts on the output.
`
`As an example, Figure 14 shows a TNY264 based 5 V, 0.5 A,
`cellular phone charger operating over a universal input range
`(85—265 VAC). The AC input is rectified and filtered by D1 —
`D4, C1 and C2 to create a high voltage DC bus connected to T1
`and in series with the high voltage MOSFET insidethe TNY264.
`The inductor (L1) forms a 1r—filter in conjuction with C1 and
`C2. The resistor R1 damps resonances in the inductor L1.
`Frequency jittering operation of TinySwitch-11 allows the use
`of a simple 11',-filter described above in combination with a
`single low value Y1-capacitor (C8) to meet worldwide
`conducted EMI standards. The addition of a shield winding in
`the transformer allows EMI to be met even with the output
`capacitively earthed. The diode D6, capacitor C3 and resistor
`R2 comprisethe clamp circuit, clamping the leakage inductance
`turn—offvoltage spike on the TinySwitch-II DRAIN pin to a safe
`Value. The secondary winding is rectified and filtered by D5
`and C5 to provide the 5 V output. Additional filtering is
`provided by L2 and C6. The output voltage is determined by the
`sum of the optocoupler U2 LED forward drop (~1 V), and
`Zener diode VR1 voltage. Resistor R8 maintains abias current
`through the Zener to ensure it is operated close to the Zener test
`current.
`
`A simple constant current circuit is implemented using the VBE
`of transistor Q1 to sense the voltage across the current sense
`
`resistor R4. When the drop across R4 exceeds the VBE of
`transistor Q1, it turns on and takes over control of the loop by
`driving the optocoupler LED. Resistor R6 provides additional
`voltage to keep the control loop in operation down to zero volts
`at the output. With the output shorted, the drop across R4 and
`R6 (~1.2 V) is sufficient to keep the Q1 and LED circuit active.
`Resistors R7 and R9 limit the forward current that could be
`
`drawn through VRl by Q1 under output short circuit conditions,
`due to the voltage drop across R4 and R6.
`
`R5 (optional) provides line under-voltage detect. At power—up
`operation is inhibited until 50 ].LA flows into the EN/UV pin.
`With a 2 MQ resistor as shown, this occurs at 100 VDC. This
`resistor also prevents the output glitching on power down.
`
`Figures 15 and 16 show examples of circuits for PC standby
`applications. They both provide two outputs, an isolated 5 V
`and a 12 V primary referenced output. The first, using TNY266P,
`provides 10 W and the second, using TNY2671’, 15 W ofoutput
`power. Both operate from an input range of 140 to 375 VDC,
`corresponding to a 230 VAC or 100/115 VAC with doubler
`input. The designs take advantage of the line under-voltage
`detect, auto—restart and higher switching frequency of
`TinySwitch-II. Operation at 132 kHz allows the use of a smaller
`and lower cost transformer core, EEl 6 for 10 W and EE22 for
`15 W. The removal of pin 6 from the 8 pin DIP TinySwitc}z
`packages provides a large creepage distance which improves
`reliability in high pollution environments such as fan cooled PC
`power supplies.
`
`Capacitor C1 provides high frequency decoupling of the high
`voltage DC supply, only necessary ifthere is a long trace length
`from the DC bulk capacitors ofthe main supply. The line sense
`resistors R2 and R3 sense the DC input voltage for line under-
`voltage. When the AC is turned off, the under—voltage detect
`feature ofthe TinySwz'lch—I1prcvcnts auto-restart glitches at the
`output caused by the slow discharge oflarge storage capacitance
`in the main converter. This is achieved by preventing the
`TinySwitch-II from switching when the input Voltage goes
`below a level needed to maintain output regulation, and keeping
`it off until the input voltage goes above the under-voltage
`threshold, when the AC is turned on again. With R2 and R3,
`giving a combined value of 2 M9, the under-voltage threshold
`is set at 200 VDC, slightly below the lowest required operating
`DC input voltage, for start-up at 170 VAC. This feature saves
`several components needed to implement the glitch-free tum-
`off compared with discrete or TOPSwitch-11 based designs.
`
`The auxiliary, primary side, winding is rectified and filtered by
`D2 and C2 to create a 12 V primary bias output voltage for the
`main power supply primary controller. In addition, via R4, this
`voltage is used to power the TinySwitch—Il. Although not
`necessary for operation, supplying the TinySwitch-II externally
`reduces the device dissipation by disabling the internal drain
`derived current source normally used to keep the BYPASS pin
`capacitor (C3) charged.
`
`The secondary winding is rectified and filtered by D3 and C6.
`For a 15 W design an additional output capacitor, C7, is required
`due to the large secondary ripple currents. The auto-restart
`function limits output current during short circuit conditions,
`removing the need to over rate D3. Switching noise filtering is
`provided by L1 and C8. The 5 V output is sensed by U2 and
`VR1. R5 is used to ensure that the Zener is biased at its test
`current.
`
`

`
`140-375 0
`
`U1
`
`TNY267P
`
`D2
`1N4148
`
`,
`
`A VR‘BZX79B3V9
`
`Figure 15.
`
`I 0 WPC Standby Supply.
`
`C5
`2.2 nF
`
`R1
`ZODKQ
`
`TNY266P
`
`A VR‘BZX79B3V9
`
`Figure 16. 15 W PC Standby Supply.
`
`TNY264-I266~268
`
`PI—2712-022601
`
`Pl-2713-022801
`
`

`
`TNY264I266-268
`
`Key Application Considerations
`
`Tinyswitch-M vs. Tinyswitch
`
`Table 2 compares the features and performance differences
`between the TNY254 device of the Tz'nySwitch family with the
`TinySwitch-II family of devices. Many of the new features
`
`eliminate the need for or reduce the cost ofcircuit components.
`Other features simplify the design and enhance performance.
`
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`
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`4Tinyswitch-flCAdvantages
`
`1
`1
`
`
`.1
`
`Switching Frequency
`and Tolerance
`Temperature Variation
`(0 - 100 °C)**
`
`+8%
`
`44 kHz i10% (@25 °C)
`
`132 kHz i6% (@25 °C)
`
`+2%
`
`14 kHz
`
`Yes — built into
`controller
`
`Single resistor
`programmable
`
`-L 7% (@25 °C)
`0%
`
`Active Frequency Jitter
`
`N/A*
`
`Transformer
`Audible Noise
`Reduction
`
`Line UV Detect
`
`N/A*
`
`Current Limit Tolerance
`Temperature Variation
`(0 — 100 °C)**
`
`i 11% (@25 °C)
`—8%
`
`Auto—Restart
`
`NIA
`
`6% effective on—time
`
`Drain Creepage at
`Package
`
`0.037” / 0.94 mm
`
`0.137” / 3.48 mm
`
`.*Not available.
`
`** See typical performance curves.
`
`Table 2. Comparison Between TinySwitch and YinySwitch-II.
`
`Design
`
`- Smaller transformerfor low cost
`- Ease of design
`- Manufacturability
`- Optimum design for lower cost
`
`- Lower EMl minimizing filter
`component costs
`
`- Practically eliminates audible noise
`with standard dip varnished
`transformer — no special construction
`or gluing required
`
`- Prevents power on/off glitches
`
`- increases power capability and
`simplifies design for high volume
`manufacturing
`
`- Limits output short—circuit current to
`less than full load current
`
`- No output diode size penalty.
`- Protects load in open loop fault
`conditions
`
`- No additional components
`required
`
`- Greater immunity to arcing as a
`result of dust, debris or other
`contaminants build—up
`
`Output Power
`Table 1 shows the practical maximum continuous output power
`levels that can be obtained under the following conditions:
`
`l. The minimum DC input voltage is 90 V or higher for 85
`VAC input, or 240 V or higher for 230 VAC input or ll5
`VAC input with a voltage doubler. This corresponds to a
`filter capacitor of3 mF/W for universal input and 1 ml-“NV
`for 230 or 115 VAC w/doubler input.
`
`2. A secondary output of 5 V with a Schottky rectifier diode.
`
`3. Assumed efficiency of 77% (TNY267 & TNY268), 75%
`(TNY266) and 73% (TNY264).
`
`4. The parts are board mounted with SOURCE pins soldered
`to sufficient area of copper to keep the die temperature at or
`below 100 °C.
`
`

`
`
`
`TNY264l266-268
`
`In addition to the thermal environment (sealed enclosure,
`ventilated, open frame, etc.,), the maximum power capability of
`TinySwitch-II in a given application depends on transformer
`core size and design (continuous or discontinuous), efficiency
`required, minimum specified input voltage,
`input storage
`capacitance, output voltage, output diode forward drop, etc.,
`and can be different from the values shown in Table I.
`
`Audible Noise
`
`The TiriySwiz‘ch-II practically eliminates any transformer audio
`noise using simple standard varnished transformer construction.
`No gluing of the cores is needed. The audio noise reduction is
`accomplished by the TinySwitch-I1 controller reducing the
`current limit in discrete steps as the load is reduced. This
`minimizes the peak flux density in the transformer when
`switching at audio frequencies.
`
`Layout
`
`Single Point Grounding
`Use a single point ground connection atthe SOURCE pin for the
`BYPASS pin capacitor and the Input Filter Capacitor (see
`Figure 17).
`
`Primary Loop Area
`The area of the primary loop that connects the input filter
`capacitor, transformer primary and TirzySwi1c/1-11 together,
`should be kept as small as possible.
`
`Primary Clamp Circuit
`A clamp is used to limit peak voltage on the DRAIN pin at turn-
`off. This can be achieved by using an RCD clamp (as shown in
`Figure 14). A Zener and diode clamp across the primary or a
`single 550 V Zener clamp from DRAIN to SOURCE can also
`be used. In all cases care should be taken to minimize the circuit
`
`path from the clamp components to the transformer and
`TinySwitch—II.
`
`Thermal Considerations
`
`Copper underneath the TinySwz’tch—I1 acts not only as a single
`point ground, but also as a heatsink. The hatched areas shown
`in Figure
`17 should be maximized for good heat sinking of
`TinySw1'tch-ll and output diode.
`
`EN/UV pin layout optimization
`The EN/UV pin connection to the optocoupler should be kept
`to an absolute minimum (less than 0.5 in.), and this connection
`should be kept away from the DRAIN pin (minimum of0.2 in.).
`If a line under-voltage detect resistor is used then the resistor
`should be mounted as close as possible to the EN/UV pin to
`minimize noise pick up.
`
`Y—Capacitor
`The placement of the Y-capacitor should be directly from the
`primary bulk capacitor positive to the common/return terminal
`on the secondary side. Such placement will maximize the EMI
`benefit of the Y—capacitor.
`
`Optocoupler
`It is important to maintain the minimum circuit path from the
`optocoupler transistor to the TinySwitch~II EN/UV and
`SOURCE pins to minimize noise coupling.
`
`Output Diode
`For best performance, the area of the loop connecting the
`secondary winding, the Output Diode and the Output Filter
`Capacitor, should be minimized. See Figure 17 for optimized
`layout. In addition, sufficient copper area should be provided
`at the anode and cathode terminals of the diode for adequately
`heatsinl-ting.
`
`Input and Output Filter Capacitors
`There are constrictions in the traces connected to the input and
`output filter capacitors. These constrictions are present for two
`reasons. The first is to force all the high frequency currents to
`flow through the capacitor (if the trace were wide then it could
`flow around the capacitor). Secondly, the constrictions minimize
`the heat transferred from the Tz'nySwitch-11 to the input filter
`capacitor and from the secondary diode to the output filter
`capacitor. The common/return (the negative output terminal in
`Figure 17) terminal of the output filter capacitor should be
`connected with a short, low resistance path to the secondary
`winding.
`In addition, the common/return output connection
`should be taken directly from the secondary winding pin andnot
`from the Y-capacitor connection point.
`
`Worst Case EMI & Efficiency Measurement
`Since identical TinySwitch-II supplies may operate at several
`different frequencies under the same load and line conditions,
`care must be taken to ensure that measurements are made under
`
`worst case conditions. When measuring efficiency or EMI
`verify