United States Patent 19
`Baumhauer, Jr. et al.
`9
`
`IIII
`USOO5506908A
`11
`Patent Numb
`5,506,908
`2
`O
`(45) Date of Patent:
`Apr. 9, 1996
`p 9
`
`(54) DIRECTIONAL MICROPHONE SYSTEM
`75l Inventors: John C. Baumhauer, Jr., Indianapolis;
`Jeffrey P. McAteer, Fishers; Alan D.
`Michel; Christopher T. Welsh, both of
`Noblesville, all of Ind., Kevin D.
`Willis, Owensboro, Ky.
`73) Assignee: AT&T Corp., Murray Hill, N.J.
`
`21 Appl. No. 268,462
`(22 Filed:
`Jun. 30, 1994
`
`Primary Examiner-Scott A. Rogers
`Assistant Examiner-Jerome Grant, I
`Attorney, Agent, or Firm-Thomas Stafford
`57
`ABSTRACT
`a
`Full directional pickup coverage is realized by employing a
`pickup arrangement which provides a plurality of audio
`polar directivity patterns, i.e., directional beams. These polar
`directivity patterns are formed in a unique embodiment of
`the invention by generating a plurality of frequency inde
`pendent time-delayed versions of a corresponding plurality
`of spatially sampled signals and by combining each of the
`plurality of spatially sampled signals with one or more
`3. GS : a
`so a wa a on a wa w
`w is a us
`as a
`ss is ss Horse Selected ones of the time delayed versions to generate at
`least a similar plurality of polar directivity patterns. More
`specifically, the spatially sampled signals are combined with
`58) Field of Search ................................. 381/155,94, 92
`the delayed versions in such a manner that a greater number
`of polar directivity patterns can be considered than the
`number of spatially sampled signals. In a specific embodi
`ment, the spatially sampled signals are acoustic (audio) and
`a plurality of microphones arranged in a predetermined
`spatial configuration is employed to obtain them.
`
`56
`
`References Cited
`U.S. PATENT DOCUMENTS
`1/1989 Elko et al. ................................ 38/54
`4,802,227
`5,193,117 3/1993 Ono et al. ................................. 38/71
`FOREIGN PATENT DOCUMENTS
`0000895 l/1989 Japan ..................................... 381/155
`
`
`
`16 Claims, 6 Drawing Sheets
`
`Page 1 of 12
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`U.S. Patent
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`Apr. 9, 1996
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`Sheet 1 of 6
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`5,506,908
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`U.S. Patent
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`Apr. 9, 1996
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`Sheet 2 of 6
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`5,506,908
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`FIG. 3
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`U.S. Patent
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`5,506,908
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`U.S. Patent
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`Apr. 9, 1996
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`Sheet 4 of 6
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`5,506,908
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`Sheet 5 of 6
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`U.S. Patent
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`Apr. 9, 1996
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`Sheet 6 of 6
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`5,506,908
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`
`
`FIG. 8
`MICROPHONE ELEMENT AYOUT AND
`CORRESPONDING POLAR RESPONSE
`
`90.0
`193
`-----------O
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`2
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`102
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`3
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`4
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`1 AND 5
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`2 AND 4
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`-Co. 8
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`5,506,908
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`1.
`DIRECTIONAL MICROPHONE SYSTEM
`CROSS REFERENCE TO RELATED
`APPLICATIONS
`U.S. patent applications Ser. No. 08/268,463 and Ser. No.
`08/258,464 were filed concurrently herewith.
`
`5
`
`TECHNICAL FIELD
`This invention relates to microphone systems and, more
`particularly, to directional microphone systems.
`
`O
`
`BACKGROUND OF THE INVENTION
`In certain audio communications systems it is desirable to
`have full room audio (acoustic) pickup. One solution to
`realize full room coverage is to use a single omni-directional
`microphone. Use of such an omni-directional microphone,
`however, has several limitations, namely, the pickup of
`sound echoes or reverberation as well as noise from the
`room. Moreover, in two-way communications systems
`using, for example, a speakerphone, the acoustic coupling
`between the receiving loudspeaker and microphone leads to
`objectionable echoes and/or annoying switching transients
`because of the required use of switched loss in the speak
`erphone.
`The limitations of the omni-directional microphone lead
`to the consideration of using directional microphones in
`such communications system. Directional gradient type
`microphone elements using internal acoustic subtraction are
`commercially available. However, use of the directional
`gradient type microphone in an apparatus requires a prior
`knowledge of the location of a talker relative to the appa
`ratus. Consequently, to obtain full room coverage, a plurality
`of such directional gradient type microphones would be
`required. This solution, however, is complex and expensive.
`
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`BRIEF DESCRIPTION OF THE DRAWINGS
`FIG. 1 is a signal flow diagram illustrating a directional
`microphone system employing one embodiment of the
`invention;
`FIG. 2 shows the spatial relationship of the microphone
`elements employed in the embodiment of FIG. 1;
`FIG. 3 shows a signal flow diagram for the balance
`network employed in the embodiments shown in FIGS. 1
`and 6;
`FIG. 4 shows in simplified form details of the voting unit
`employed in the embodiment of FIG. 1;
`FIG. S shows polar directivity patterns for the configu
`ration of microphone elements shown in FIG. 2 resulting
`from employing the embodiment of FIG. 1;
`FIG. 5A illustrates cardioid and hypercardioid polar direc
`tivity patterns;
`FIG. 6 is a signal flow diagram illustrating a directional
`microphone system employing another embodiment of the
`invention;
`FIG. 7 shows the spatial relationship of the microphone
`elements employed in the embodiment of FIG. 6; and
`FIG. 8 shows polar directivity patterns for the configu
`ration of microphone elements shown in FIG. 7 resulting
`from employing the embodiment of FIG. 6.
`
`DETAILED DESCRIPTION
`FIG. 1 illustrates in simplified form a signal flow diagram
`for signal channels associated with three microphone ele
`ments employing one embodiment of the invention. It is
`noted that the signal flow diagram of FIG. 1 illustrates the
`signal flow processing algorithm which may be employed in
`a digital signal processor (DSP) to realize the invention. It
`is noted, however, although the preferred embodiment of the
`invention is to implement it on such a digital signal proces
`sor, that the invention may also be implemented as an
`integrated circuit or the like. Such digital signal processors
`are commercially available, for example, the DSP 1600
`family of processors available from AT&T.
`Shown in FIG. 1 are microphone elements 101, 102 and
`103, which in this embodiment, are arranged in an equilat
`eral triangle as shown in FIG. 2. As shown in FIG. 2,
`microphone elements 101,102 and 103 are placed at the
`vertices of the equilateral triangle with a predetermined
`spacing "d" between the vertices. In this example, the
`spacing d between the vertices is approximately 0.85 inches.
`An output signal from microphone element 101 is supplied
`via amplifier 104 and Codec 105 to DSP 106 and therein to
`balance network 107. DSP 106 includes the digital signal
`flow processing to realize the invention. Also shown is
`microphone element 102 whose output is supplied via
`amplifier 108 and Codec 109 to DSP 106 and therein to
`balance network 107. Finally, an output signal from micro
`phone element 103 is supplied via amplifier 110 and Codec
`111 to DSP 106 and therein to balance network 107. In one
`example, employing the invention, microphone elements
`101, 102 and 103 are so-called omni-directional micro
`phones of the well-known electret-type. Although other
`types of microphone elements may be utilized in the inven
`tion, it is the electret type that are the preferred ones because
`of their low cost. Codecs 105, 109 and 111 are also well
`known in the art. One example of a Codec that can advan
`tageously be employed in the invention is the T7513B
`Codec, also commercially available from AT&T. In this
`example, the digital signal outputs from Codecs 105, 109
`
`40
`
`SUMMARY OF THE INVENTION
`Full directional pickup coverage is realized by employing
`a pickup arrangement which provides a plurality of polar
`directivity patterns, i.e., a plurality of directional beams.
`These polar directivity patterns are formed in a unique
`embodiment of the invention by generating a plurality of
`frequency independent time-delayed versions of a corre
`sponding plurality of spatially sampled signals and by
`45
`combining each of the plurality of spatially sampled signals
`with one or more selected ones of the time delayed versions
`to generate at least a similar plurality of polar directivity
`patterns. More specifically, the spatially sampled signals are
`combined with the delayed versions in such a manner that a
`greater number of polar directivity patterns can be consid
`ered than the number of spatially sampled signals.
`In another embodiment, the spatially sampled signals are
`also combined with each other in such a manner to form
`additional polar directivity patterns.
`In a specific embodiment, the spatially sampled signals
`are acoustic (audio) and a plurality of microphones arranged
`in a predetermined spatial configuration is employed to
`obtain them.
`A technical advantage of the invention is that the number
`of polar directivity patterns generated to handle the full
`directional, e.g., room, coverage pickup is greater than the
`number of microphone inputs required. Another technical
`advantage is the ability to alter the shape of the audio polar
`directivity patterns solely through changing the software
`code.
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`and 111 are encoded in the well-known mu-law PCM
`format, which in DSP 106 must be converted into a linear
`PCM format. This mu-law-to-linear PCM conversion is well
`known. Balance network 107 is employed to balance, i.e.,
`match, the long term average broad band gain of the signal
`channels associated with microphone elements 101, 102 and
`103 to one another. In this example, the long term average
`broad band gain of the signal channels associated with
`microphone elements 101 and 103 are balanced to the signal
`channel associated with microphone element 102. Details of
`balance network 107 are shown in FIG. 3 and described
`below.
`More specifically, DSP 106 first forms a plurality of polar
`directivity patterns to provide full pick up coverage of a
`particular space, for example, a room, stage, arena, area or
`the like and then vote on the polar directivity pattern (or
`patterns) that has the best signal-to-noise ratio, thus picking
`up the desired signal source. In this example, the polar
`directivity patterns are acoustic (audio) and are in predeter
`mined spatial orientation relative to each other in order to
`provide full 360° coverage of the particular space. To this
`end the balanced microphone signal channel outputs A, B
`and C corresponding to microphones 101,102 and 103,
`respectively, from balance network 107 are delayed by delay
`units 112,113 and 114, respectively. In this example, each of
`delay units 112, 113 and 114 provides a time delay interval
`equivalent to the time that sound takes to travel the distance
`d from one of the microphone pick up locations to another
`to yield frequency independent time delayed versions A, B
`and C respectively. The delayed signal outputs A, B' and C
`from delay units 112, 113 and 114 are then algebraically
`combined with the non-delayed versions A, B and C, respec
`tively, from balance network 107 via algebraic summing
`units 121 through 126 to generate six signals representing
`cardioid polar directivity patterns. Alternatively, for distance
`d being twice the above noted value, and the time delay
`interval being equivalent to one-third the time it takes sound
`to travel the new distance, hypercardioid polar directivity
`patterns will be generated for the six polar directivity
`patterns. FIG. 5A shows the relationship of a cardioid polar
`directivity pattern (solid outline) and a hypercardioid polar
`directivity pattern (dashed outline). Note that by further
`changing the delay interval of each of delay units 112, 113
`and 114 and/or the spacing "d', the resulting polar directiv
`ity patterns can be changed, as desired. Changing this delay
`45
`interval is readily realized simply by reprogramming DSP
`106.
`FIG. 5 illustrates the relationship of the equilateral tri
`angle configuration of microphones 101, 102 and 103 and
`the resulting six cardioid polar directivity patterns, as well
`as, the resulting three "FIG. 8' polar directivity patterns
`which will be discussed below. The six cardioid polar
`directivity patterns result from the algebraic summing of the
`delayed versions of the balanced channel signals A, B' and
`C with the non-delayed balanced channel signals A, B and
`C, respectively. Thus, summing unit 121 yields at circuit
`point 131 a signal (B-A) representative of a cardioid polar
`directivity pattern having its null in the direction of micro
`phone 101 and having its maximum sensitivity in the
`direction of microphone 102 (shown in dashed outline in
`FIG. 5 from direction 2 to direction 5). Summing unit 122
`provides at circuit point 132 a signal (C-A) representative of
`a cardioid polar directivity pattern having its null also in the
`direction of microphone 101 and having its maximum
`sensitivity in the direction of microphone 103 (shown in
`65
`dashed outline in FIG. 5 from direction 3 to direction 6).
`Summing unit 123 yields at circuit point 133 a signal (A-B)
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`representative of a cardioid polar directivity pattern having
`its null in the direction of microphone 102 and having its
`maximum sensitivity in the direction of microphone 101
`(shown in solid outline in FIG. 5 from direction 5 to
`direction 2). Summing unit 124 yields at circuit point 134 a
`signal (C-B") representative of a cardioid polar directivity
`pattern having its null in the direction of microphone 102
`and having its maximum sensitivity in the direction of
`microphone 103 (shown in solid outline in FIG. 5 from
`direction 4 to direction 1). Summing unit 125 yields at
`circuit point 135 a signal (A-C) representative of a cardioid
`polar directivity pattern having its null in the direction of
`microphone 103 and having its maximum sensitivity in the
`direction of microphone 101 (shown in solid outline in FIG.
`5 from direction 6 to direction 3). Summing unit 126 yields
`at circuit point 136 a signal (B-C) representative of a
`cardioid polar directivity pattern having its null in the
`direction of microphone 103 and having its maximum
`sensitivity in the direction of microphone 102 (shown in
`dashed outline in FIG. 5 from direction 1 to direction 4). The
`signals at circuit points 131 through 136, representative of
`the cardioid polar directivity patterns, are supplied to voting
`unit 140 and to multiplier units 141 through 146, respec
`tively. The purpose of the cardioid polar directivity patterns
`generated by summing units 121 through 126 is to pick up
`single acoustic sources, for example, single talkers. In this
`example, the six cardioid polar directivity patterns are
`pointing in predetermined fixed directions and are spaced
`60° apart from each other. Algebraic summing units 127,
`128 and 129 are employed to derive so-called FIG. 8 polar
`directivity patterns capable of picking up acoustic sources
`on opposite sides of the pickup system which are operating
`simultaneously, for example, two simultaneous talkers.
`Summing unit 127 provides a signal (A-B) at circuit point
`137 representative of a FIG. 8 polar directivity pattern that
`is sensitive, in this example, to talkers at the ends of a
`directional line passing through microphones 101 and
`microphone 102 (shown in FIG.5 as a FIG. 8 for directions
`2 and 5). Summing unit 128 provides a signal (B-C) at
`circuit point 138 representative of a FIG.8 polar directivity
`pattern that picks up, in this example, talkers at the ends of
`a directional line passing through microphone 102 and
`microphone 103 (shown in FIG. 5 as a FIG. 8 for directions
`1 and 4). Summing unit 129 provides a signal (A-C)
`representative at circuit point 139 of a FIG. 8 polar direc
`tivity pattern that picks up, in this example, talkers at the
`ends of a directional line passing through microphone 101
`and microphone 103 (shown in FIG. 5 as a FIG. 8 for
`directions 3 and 6). The signals at circuit points 137,138 and
`139 are also supplied to voting unit 140 and to multiplier
`units 147,148 and 149, respectively.
`Voting unit 140 determines the optimum weighting pro
`vided by each of the signal channels 131 through 139 at
`outputs 151 through 159, respectively. Details of voting unit
`140 are shown in FIG. 4 and described below. The signals
`representative of these weightings from outputs 151 through
`159 are also supplied to multipliers 141 through 149 respec
`tively, to weight each channel in accordance with its desir
`ability to be represented in the output. Algebraic summing
`unit 160 algebraically combines the weighted output signals
`from each of multipliers 141 through 149. Then, Codec 161
`converts the summed output signal into an analog form. The
`output of Codec 161 is then transmitted as desired.
`FIG. 3 shows in simplified form a signal diagram illus
`trating the operation of balance network 107. The mu-law
`PCM output from each of Codecs 105, 109 and 111 is
`converted to linear PCM format (not shown) in DSP 106.
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`5
`Then, the linear PCM representations of the outputs from
`Codec 105 and Codec 111 are supplied to gain differential
`correction factor generation units 301 and 302, respectively.
`Because the long term average broad band gain of the
`microphone signal channels corresponding to microphones
`101 and 103 are being matched to the signal channel of
`microphone 102, in this example, the linear PCM format
`output of Codec 109 does not need to be adjusted. Since each
`of gain differential correction factor generation units 301
`and 302 is identical and operates the same, only gain
`differential correction factor generation unit 301 will be
`described in detail. To this end, the elements of each of gain
`differential correction factor generation units 301 and 302
`have been labeled with identical numbers.
`The matching, i.e., balancing, of the long term average
`broad band gain of the signal channels corresponding to
`microphone elements 101 and 102 is realized by matching
`the signal channel level corresponding to microphone ele
`ment 101 to that of microphone element 102. To this end, the
`linear PCM versions of the signal from Codec 105 is
`supplied to multiplier 303. Multiplier 303 employs again
`differential correction factor 315 to adjust the gain of the
`linear PCM version of the signal from Codec 105 to obtain
`an adjusted output signal 316, i.e., A, for microphone 101.
`As indicated above, the linear PCM version of the signal
`from Codec 109 does not need to be adjusted and this signal
`is output B from balance network 107. The adjusted output
`Cofbalance network 107 is from gain differential correction
`factor generation unit 302.
`The gain differential correction factor 315 is generated in
`the following manner: adjusted microphone output signal
`316 is squared via multiplier 304 to generate an energy
`estimate value 305. Likewise, the linear PCM version of the
`output signal from Codec 109 is squared via multiplier 307
`to generate energy estimate value 308. Energy estimate
`values 305 and 308 are algebraically subtracted from one
`another via algebraic summing unit 306, thereby obtaining
`a difference value 309. The sign of the difference value 309
`is obtained using the signum function 310, in well known
`fashion, to obtain signal 311. Signal 311 will be either minus
`40
`one (-1) or plus one (+1) indicating which microphone
`signal channel had the highest instantaneous energy. Minus
`one (-1) represents microphone 101, and plus one (+1)
`represents microphone 102. Multiplier 312 multiplies signal
`311 by a constant K to yield signal 313 which is a scaled
`version of signal 311. In one example, not to be construed as
`limiting the scope of the invention, K typically would have
`a value of 10 for a 22.5 ks/s (kilosample per second)
`sampling rate. Integrator 314 integrates signal 313 to pro
`vide the current gain differential correction factor 315. The
`integration is simply the sum of all past values. In another
`example, constant K would have a value of 5x10 for an 8
`ks/s sampling rate. Value K is the so-called "slew" rate of
`integrator 314.
`FIG. 4 shows, in simplified block diagram form, details of
`voting unit 140. Specifically, shown are so-called talker
`signal-to-noise estimation units 401 through 409. It is noted
`that each of talker signal-to-noise ratio estimate units 401
`through 409 are identical to each other, Consequently, only
`talker signal-to-noise ratio estimation unit 401 will be
`described in detail. A signal representative of the cardioid
`polar directivity pattern generated by summing unit 121 is
`supplied via 131 to talker signal-to-noise ratio estimation
`unit 401 and therein to absolute value generator unit 410.
`The absolute value of the signal supplied via 131 is obtained
`and is then applied to peak detector 411 in order to obtain its
`peak value over a predetermined window interval, in this
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`example, 8 ms. The obtained peak value is supplied to
`decimation unit 412 which obtains the generated peak value
`every 8 ms, in this example, clearing the peak detector 411
`and supplies the obtained peak value to short term filter 413
`and long term filter 414. Filters 413 and 414 provide noise
`guarding of signals from stationary noise sources. Short
`term filter 413, in this example, is a non-linear first order low
`pass filter having a predetermined rise time constant, for
`example, of 8 ms and a fall time, for example, of 800 ms.
`The purpose of filter 413 is to generally follow the envelope
`of the detected wave form. Long term filter 414 is also a
`non-linear first order low pass filter having, in this example,
`arise time of 8 seconds and a fall time of 80 ms. The purpose
`of filter 414 is to track the level of background interference.
`Ten times the logarithm of the filtered output signal from
`short term filter 413 is obtained via logarithm (LOG) unit
`415 and supplied to one input of algebraic summing unit
`417. Similarly, ten times the logarithm of the filtered output
`signal from long term filter 414 is obtained via LOG unit 416
`and supplied to another input of algebraic summing unit 417.
`The LOG values from LOG units 415 and 416 are algebra
`ically subtracted in algebraic summing unit 417. The result
`ing difference signal is supplied to maximum (MAX) detec
`tor 418. Similarly, the outputs from talker signal-to-noise
`estimation units 402 through 409 are also supplied to MAX
`detector 418. MAX detector 418 provides a true output, i.e.,
`a logical 1, for the corresponding talker signal-to-noise
`estimation unit output having the largest value output during
`the sampling window, in this example, 8 ms. MAX detector
`418 also provides a false, i.e., logical 0, output for the signal
`channels corresponding to the other talker signal-to-noise
`estimation units. Additionally, MAX detector 418 provides
`an output only when a difference between the logarithm of
`the maximum signal-to-noise ratio value minus the loga
`rithm of the minimum signal-to-noise ratio value obtained
`during the 8 ms window is greater than a predetermined
`value, in this example, 3 dB, and when the logarithm of the
`maximum signal-to-noise ratio value is greater than a second
`predetermined value, in this example, 15 dB. The outputs
`from MAX detector 418 are supplied to up/down (UID)
`counters 421 through 429. Each of U/D counters 421
`through 429 increase their count value by a predetermined
`value, in this example, 0.05, each time the signal supplied
`from MAX detector 418 is true up to a predetermined
`maximum value of, in this example, one (1). Likewise, if the
`signal supplied from MAX detector 418 to U/D counters 421
`through 429 is false, the counters count down by the
`predetermined value of, in this example, 0.05 to another
`predetermined value of, in this example, zero (0). Each of
`counters 421 through 429 count either up or down once
`every window interval of 8 ms, in this example. When the
`above noted conditions regarding the values of the logarithm
`of the maximum and minimum signal-to-noise ratios are not
`met, all of counters 421 through 429 maintain their present
`count. The outputs from U/D counters 421 through 429 are
`the outputs 151 through 159, respectively, of voting unit
`140.
`FIG. 6 illustrates, in simplified form, a flow diagram for
`signal channels associated with microphone elements 101,
`102 and 103 employing another embodiment of the inven
`tion. The spatial configuration of microphone elements 101,
`102 and 103 in this embodiment, includes two legs extend
`ing from a single point at a right angle and having one of the
`microphones at each end of the legs and at the single point.
`Thus, as shown in FIG.7 microphone element 101 is at one
`end of one of the legs, microphone element 102 is at the
`single point and microphone element 103 is at the end of the
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`other leg of the right angle. As shown in FIG. 7, the spacing
`between the microphones is "d'. It is noted that the signal
`flow diagram of FIG. 6 employs some of the elements of the
`signal flow diagram shown in FIG. 1. The elements which
`are similar have been similarly numbered and since their
`operation is identical to that of FIG. 1 they will not be
`described again in detail. It is noted, however, that instead of
`employing nine summing units, six of which generated the
`cardioid polar directivity patterns and three of which gen
`erated the FIG. 8 polar directivity patterns in the embodi
`ment of FIG.1, the embodiment of FIG. 6 employs algebraic
`summing units 121, 123, 124 and 126 to generate four
`cardioid polar directivity patterns and algebraic summing
`units 127 and 128 to generate two FIG. 8 polar directivity
`patterns. Voting unit 140 generates the weighted signal-to
`noise ratio values only for the signals supplied at circuit
`points 131, 133,134,136, 137 and 138 from their associated
`algebraic summing units. Thus, only six signal channels are
`being voted on and similarly only those six signal channels
`are being weighted via multipliers 141, 143, 144, 146, 147
`and 148 via weighted outputs 151,153, 154, 156, 157 and
`158, respectively, from voting unit 140. Algebraic summing
`unit 160 algebraically sums the weighted outputs from
`multipliers from 141, 143, 144, 146, 147 and 148 to obtain
`the desired digital output. This digital output is supplied to
`Codec 161 which converts it to audio form for further
`transmission as desired.
`FIG. 8 illustrates the relationship of the right triangle
`configuration of microphones 101, 102 and 103 and the
`resulting four cardioid polar directivity patterns as well as
`the resulting two FIG. 8 polar directivity patterns. The four
`cardioid polar directivity patterns result from the algebraic
`summing of the delayed versions of the balanced channel
`signals, A, B' and C with the non-delayed balanced channel
`signals A, B and C, respectively. Thus, summing unit 121
`yields, at circuit point 131, a signal (B-A) representative of
`a cardioid polar directivity pattern having its null in the
`direction of microphone 101 and having its maximum
`sensitivity in the direction of microphone 102 (shown in
`FIG. 8 from direction 2 to direction 4). Summing unit 123
`provides, at circuit point 133, a signal (A-B) representative
`of a cardioid polar directivity pattern having its null in the
`direction of microphone 102 and having its maximum
`sensitivity in the direction of microphone 101 (shown in
`FIG. 8 from direction 4 to direction 2). Summing unit 124
`yields, at circuit point 134, a signal (C-B) representative of
`a cardioid polar directivity pattern having its null also in the
`direction of microphone 102 and having its maximum
`sensitivity in the direction of microphone 103 (shown in
`FIG. 8 from direction 3 to direction 1). Summing unit 126
`yields, at circuit point 136, a signal (B-C) representative of
`a cardioid polar directivity pattern having its null in the
`direction of microphone 103 and having its maximum
`sensitivity in the direction of microphone 102 (shown in
`FIG. 8 from direction 1 to direction 3). Again, the signals at
`circuit points 131, 133, 134 and 136 are supplied to voting
`unit 140 and to multiplier units 141,143, 144 and 146,
`respectively. The purpose of the cardioid polar directivity
`patterns generated by summing units 121,123, 124 and 126
`is also to pick up single acoustic sources. Algebraic sum
`60
`ming units 127 and 128 are employed to derive so-called
`FIG. 8 polar directivity patterns capable of picking up
`acoustic sources on opposite sides of the pick up system
`which are operating simultaneously, for example, two simul
`taneous talkers. Summing unit 127 provides a signal (A-B)
`at circuit point 137 representative of a FIG. 8 polar direc
`tivity pattern that is sensitive, in this example, to talkers at
`
`8
`the ends of a directional line passing through microphones
`101 and 102 shown in FIG. 8 as a FIG. 8 for directions 2 and
`4. Summing unit 128 provides a signal (B-C) at circuit point
`138 representative of a FIG. 8 polar directivity pattern that
`picks up, in this example, talkers at the ends of a directional
`line passing through microphone 102 and microphone 103
`shown in FIG. 8 as a FIG. 8 for directions 1 and 3.
`Although the embodiments of the invention have been
`described in the context of picking up acoustic (audio)
`signals, it will be apparent to those skilled in the art that the
`invention can also be employed to pick up other energy
`sources; for example, those which radiate radio frequency
`waves, ultrasonic waves, or other acoustic waves in liquids
`and solids or the like.
`What is claimed:
`1. A directional pickup system comprising:
`a plurality of means for generating frequency independent
`time-delayed versions of a corresponding plurality of
`spatially sampled signals; and
`means for combining each of the plurality of spatially
`sampled signals with one or more predetermined ones
`of the time delayed versions to generate representations
`of at least a similar plurality of polar directivity pat
`terns, said means for combining including means for
`combining each of the plurality of spatially sampled
`signals with selected ones of the time delayed versions
`to generate a number of polar directivity patterns which
`is greater than said plurality of spatially sampled sig
`nals.
`2. The system as defined in claim 1 wherein said means
`for generating includes means for selecting delay intervals
`based on prescribed criteria for a particular polar directivity
`pattern.
`3. The system as defined in claim 1 wherein said means
`for combining includes means for algebraically subtracting
`each of the plurality of spatially sampled signals from
`selected ones of the time delayed versions.
`4. The system as defined in claim 1 further including
`means supplied with said representations of said plurality of
`polar directivity patterns and being responsive thereto to
`select the polar directivity pattern that has the highest
`estimated signal-to-background noise ratio with regard to a
`desired signal source.
`5. The system as defined in claim 1 further including
`means supplied with said plurality of spatially sampled
`signals for substantially matching the long term average
`broad band gain of signal channels associated with said
`spatially sampled signals to one another.
`6. The system as defined in claim 1 wherein at least two
`polar directivity patterns are generated, each of said polar
`directivity patterns having a prescribed width and direction
`that is selected to cover a predetermined area of interest.
`7. The system as defined in claim 1 wherein the plurality
`of polar directivity patterns is six be